Dual-Channel Colorimeter with Programmable Gain Transimpedance Amplifier and Synchronous Detector

　　Circuit Function and Advantages

　　The circuit shown in Figure 1 is a two-channel colorimeter that integrates a modulated light source transmitter and a synchronous detector receiver. The circuit measures the ratio of light absorbed by a sample and a reference container at three different wavelengths.

　　This circuit provides an effective solution for many chemical analysis and environmental monitoring instruments used to measure concentrations and characterize materials by absorption spectroscopy.

　　The photodiode receiver conditioning path includes a programmable gain transimpedance amplifier to convert the diode current to a voltage and allow analysis of different liquids with widely varying light absorption. A 16-bit Σ-Δ analog-to-digital converter (ADC) provides additional dynamic range, ensuring adequate resolution over a wide range of photodiode output currents.

　　Using a modulated light source and synchronous detector rather than a constant current (DC) source eliminates measurement errors caused by ambient light and low frequency noise and provides greater accuracy.

　　Figure 1. Dual-Channel Colorimeter with Programmable Gain Transimpedance Amplifier and Lock-In Amplifier (Simplified Schematic: All Connections and Decoupling Not Shown)

　　Circuit Description

　　The AD8618 quad op amp forms three simple current sources to drive the LEDs with constant current. The EVAL-SDP-CB1Z generates a 5 kHz clock that modulates one LED through the ADG633 single-pole double-throw (SPDT) switch to turn the reference voltage of its current source on or off. The current sources of the other two LEDs are set to 0 V to turn them off when not in use.

　　A beam splitter sends half of the light through the sample container and the other half through the reference container. Depending on the type and concentration of the medium in each container, the containers absorb different amounts of light. A photodiode on the other side of each container generates a small amount of current proportional to the amount of light received.

　　The first stage of each channel consists of an AD8615 op amp configured as a transimpedance amplifier to convert the photodiode output current to a voltage. The AD8615 is a good choice as a photodiode amplifier because it has very low input bias current (1 pA), input offset voltage (100 μV), and noise (8 nV/√Hz). Although the signal is subsequently ac-coupled, it is still important to minimize dc errors in this stage to avoid loss of dynamic range. The op amp input bias current is multiplied by the feedback resistor value at the output as the offset voltage. The op amp input offset voltage at the output with gain is determined by the feedback resistor and the photodiode shunt resistance. In addition, any op amp input voltage offset across the photodiode will result in an increase in the photodiode dark current.

　　Figure 2 shows a typical transconductance amplifier with a single feedback resistor and its ideal transfer function.

　　Figure 2. Transconductance amplifier transfer function.

　　Because some solutions under test can have very strong absorption characteristics, it is sometimes necessary to use a large feedback resistor to measure the very small current produced by the photodiode while being able to measure the large currents corresponding to highly dilute solutions. To address this challenge, the photodiode amplifier in Figure 1 has two different selectable gains. One gain is set to 33 kΩ and the other is set to 1 MΩ. When a single SPDT switch is connected to the output of the op amp to switch the feedback resistor, the on resistance of the ADG633 can cause a transimpedance gain error.

　　To avoid this problem, a better configuration is shown in Figure 3, where the ADG633 inside the feedback loop selects the desired resistance, while a second switch connects the next stage of the system to the selected feedback loop. The voltage at the output of the amplifier is:

　　VTIA OUTPUT = IPHOTODIODE × RFEEDBACK

　　Rather than

　　VTIA OUTPUT = IPHOTODIODE × (RFEEDBACK + RON ADG633) With the ADG633 outside the feedback loop, the output impedance of this stage is simply the on resistance of the ADG633 (typically 52 Ω) rather than the very low output impedance associated with the output of the op amp when operating in closed loop.

　　Note that for stability reasons, feedback capacitor CFx is required to compensate for the total input capacitance (diode capacitance plus op amp input capacitance) and the pole created by feedback resistor RFx. For more details on this analysis, see Part 5 of Practical Design Tips for Sensor Signal Conditioning.

　　Even the best rail-to-rail output amplifiers, such as the AD8615, cannot swing their outputs completely to the rails. In addition, the input offset voltage on the AD8615 can be negative, albeit very small. Rather than using a negative supply to ensure that the amplifier is not clipped, the ADR4525 reference biases the photodiode and amplifier to 2.5 V, which can be driven to 0 V. The analog and digital sections of the board are powered by 5 V linear regulators.

　　Figure 3. Programmable gain transconductance amplifier.

　　The photodiode amplifier output voltage can swing from 2.5 V to 5.0 V. For the 33 kΩ range, the 2.5 V output range corresponds to a full-scale photodiode current value of 75.8 μA. For the 1 MΩ range, this corresponds to 2.5 μA. When operating with the 1 MΩ gain setting, it is important to protect the photodiode from external light to prevent the amplifier from saturating. Although the synchronous rectifier described below will greatly attenuate any frequency that is not synchronized with the LED clock, it will not function properly if the previous stage is attenuated. The gain setting for each channel can be independently selected using the EVAL-SDP-CB1Z board.

　　The next stage is a simple buffered ac-coupled filter. The filter cutoff frequency is set to 7.2 Hz; it removes any output offset voltage and attenuates low-frequency light pollution from incandescent and fluorescent lamps and any other stray light entering the photodiode. At the same time, the output of the ADR4525 also biases the circuit to 2.5 V; therefore, the output signal swing of this stage is nominally between 1.25 V and 3.75 V.

　　Following the ac-coupled filter is a synchronous rectifier circuit that uses the AD8271 difference amplifier and the ADG733 triple SPDT switch. The ADG733 internal switches are connected in series with the AD8271's internal 10 kΩ gain-setting resistors; therefore, the ADG733's 4.5 Ω maximum on-resistance contributes only 0.05% gain error, and the temperature drift is less than 1 ppm/°C.

　　The rest of the system uses ADG633 switches because of their very low leakage current and low parasitic capacitance.

　　When the clock driving the LED is in the high state, the switches within the ADG733 configure the AD8271 according to the following simple transfer function:

　　VO = VIN

　　in:

　　VO is the output of the synchronous detector.

　　VIN is the input to the synchronous detector and can range from 2.5 V to 3.75 V.

　　In this configuration, the synchronous rectifier acts as a unity-gain amplifier.

　　When the clock driving the LED is in the low state, the switches within the ADG733 configure the AD8271 according to the following transfer function:

　　VO = 2VREF − VIN

　　in:

　　VREF is the 2.5 V output of the ADR4525.

　　The VIN range is 1.25 V to 2.5 V.

　　In this case, when the input is 1.25 V (the minimum voltage that the ac-coupled stage can output), the output of the synchronous rectifier is 3.75 V, and when the input is 2.5 V (the middle of the ac-coupled stage), the output of the synchronous rectifier is 2.5 V. In this configuration, the synchronous rectifier has a gain of −1 and is biased around the +2.5 V reference voltage.

　　Figure 4. System block diagram and time domain waveforms for each step.

　　Figure 4 shows the system block diagram and the voltage range of each stage. The result of the synchronous rectification circuit is a variable DC voltage that can vary from 2.5 V (no light reaching the photodiode) to 3.75 V (full-scale light input). This output voltage corresponds to a full-scale output swing of 1.25 V.

　　This circuit filters signals that are not synchronous with the LED clock (or odd harmonics since the clock waveform is a square wave). In the frequency domain, the low-pass filter at the output of the AD8271 looks like a band-pass filter around the LED clock frequency. The lower the bandwidth of this filter, the better the synchronous rectifier can reject out-of-band noise. The filter cutoff frequency is set to 16 Hz due to the trade-off between noise rejection and settling time. It is important to note that the filter bandwidth is approximately equal to the LED clock. For example, if the LED modulation is 5 kHz, the 3 dB passband range of the synchronous detector is 4.984 kHz to 5.016 kHz.

　　The final stage of the system is the low noise, 16-bit, Σ-Δ ADC AD7798. This ADC integrates a built-in programmable gain amplifier (PGA) with differential inputs. A 2.5 V reference is connected to the AIN pin, and the PGA gain is set to 2 to allow it to map the 2.5 V to 3.75 V output of the synchronous rectifier to a full-scale 16-bit output. In addition, the output filter of the AD7798 provides a minimum of 65 dB rejection at 50 Hz and 60 Hz, further attenuating any noise from the synchronous detector.

　　To verify that the front-end circuitry does not contribute excessive noise to the system, data was acquired with the LEDs disabled. The synchronous detector still operates at the LED clock frequency, but does not detect any optical signals that are synchronous to that clock. Therefore, it removes all dc and ac signals except for the errors generated by the AD8271 and the ADC. Figure 5 shows the noise in this configuration, which is less than 1 LSB for one channel (the ADC input is centered between two codes) and 1 LSB peak-to-peak for the other channel (the ADC input is in the transition region between two adjacent codes). Also, note that the measured voltage is negative, at a few mV, which is expected performance given the typical offset error distribution of the AD8271.

　　Figure 5. ADC voltage with LED source disabled

　　Common changes

　　Changing the value of the feedback resistor on the photodiode amplifier changes the amplifier gain. This is an easy way to customize the circuit for a specific application at different light levels. However, the compensation capacitor must also be changed to maintain the bandwidth and ensure the stability of the amplifier.

　　For very low light level measurement systems, the synchronous detector's low-pass filter cutoff frequency can be set to a much lower frequency value to provide the best performance, but at the expense of a longer measurement period.

　　Since the light output of the LED changes with temperature, the system measures the ratio of the sample and reference channels. The gain tolerance of the photodiode is ±11% maximum; therefore, there is some drift in the ratio as the LED output changes with time and temperature. By adding an optical feedback loop to control the amplitude, the LED can significantly reduce the degree of light change with temperature, even making single-channel accurate measurement possible. Figure 7 shows the ratio of the reference channel to the sample channel readings during a typical 200-sample acquisition period.

　　Figure 6. Calibrated scale reading (red LED on, distilled water in sample and reference containers)