Analysis of Equivalent Parallel Capacitance of Planar Magnetic Integrated EMI Filter

Electromagnetic interference problems are often complicated by subtle coupling paths in circuits , and their clear and effective solutions generally rely on engineers' experience or numerical simulations based on classical models. Engineers are happy that if all non-contact electromagnetic interference can be modeled with traditional lumped elements, and this model can be combined with conversion circuit diagrams to depict all conducted interference and coupled interference. This makes analysis and prediction easier.

The lumped element circuit model is suitable for analyzing and predicting electromagnetic interference in the frequency band of 0 to 30 MHz. In many previous studies and the process of deriving so-called simple models for analysis, it is usually crucial to understand the important paths. However, these parameters are difficult to obtain in the case where there may be tiny coupling paths. For this reason, this paper uses a common lumped circuit model that can be used to analyze all non-contact electromagnetic interference that is prone to occur to accurately analyze the planar PCB EMI integrated filter. Because the model is closer to the actual situation, it can more truthfully reflect the filtering performance of the planar PCB EMI integrated filter.

Traditional discrete components are often considered as ideal devices (i.e. pure resistors , pure inductors, pure capacitors ) at low frequencies. However, at higher frequencies, the device characteristics will deviate far from their ideal characteristics. Therefore, the influence of high-frequency distributed parameters on the device characteristics must be considered. The main components in EMI power filters are inductors and capacitors . Therefore, this article mainly discusses the influence of high-frequency distributed parameters of inductors and capacitors on their filtering performance.

Because EMI filters are mainly used to filter out high-frequency noise and high-frequency interference signals, and because the equivalent parasitic parameters of capacitors and inductors will seriously affect the high-frequency performance of power filters, improving the high-frequency performance of EMI power filters should focus on improving the high loss of EMI filters, reducing the equivalent parallel capacitance of common-mode inductors, and reducing the equivalent series inductance of capacitors. In order to achieve the above purpose, this paper proposes a planar magnetic integrated structure for EMI power filters.

1 Planar magnetic integrated structure of EMI power filter

In order to realize the planar magnetic integrated structure of the EMI power filter, an LC planar magnetic integrated structure as shown in FIG1 is introduced here.

The structure in Figure 1 is formed by directly embedding two winding conductors on both sides of a flat insulating plate. It controls the connection points of the windings so that A and D are input points and C and D are output points. Thus, a low-pass filter as shown in Figure 2 can be obtained.

1.1 Implementation of integrated CM (common mode) filter

Under common mode excitation, the EMI filter can be equivalent to two parallel low-pass filters. Therefore, the integrated CM filter can be realized by two integrated LC winding coils as shown in Figure 3. The two integrated LC winding coils in Figure 3 are connected to form a low-pass filter, and there is a strong magnetic coupling between them. The corresponding equivalent circuit is shown in Figure 4.

1.2 Implementation of integrated DM (differential mode) filter

The equivalent circuit of the differential mode filter is a collapse-type low-pass filter, and its filter inductance value is very small. It is about in the range of 10 to 20 μH. The two filter capacitors have larger values, and their values ​​are in the range of 0.1 to 1 μF. Similar to the discrete EMI filter, in the integrated EMI filter, the differential mode inductance is also realized by using the leakage inductance of the integrated CM choke. In terms of controlling the leakage inductance value, the planar CM choke has more flexibility, and it can insert an additional layer of magnetic material between the two winding coils. Therefore, there is no need to change the number of turns of the CM inductor, so that the leakage inductance value can be changed by adjusting the magnetic permeability and effective area of ​​the magnetic material. This provides an opportunity for the decoupling of DM and CM inductors. Figure 5 shows the integrated filter differential mode inductor model, and Figure 6 shows the equivalent circuit diagram of the integrated filter differential mode inductor.

The DM capacitor can be realized by another integrated LC winding coil connected as a capacitor. It has only one turn or less than one turn. Figure 7 and Figure 8 show its differential mode capacitance and its equivalent circuit respectively.

Figures 9 and 10 are the two-dimensional and three-dimensional structures of the planar PCB EMI integrated filter, respectively. All parameters of the filter are obtained through simulation of the model.

2. Filter material selection considerations

It is much easier for power supply filters to suppress high-frequency EMI signals than to eliminate low-frequency EMI signals. Usually, the differential mode inductance formed by the leakage inductance L of the common mode choke can eliminate the conducted interference level of 0.3 to 30 MHz. The design and selection of filter inductors must be based on the actual needs of the circuit. Generally, the differential mode interference plays a leading role in the range of 0.01 to 0.1 MHz. The differential mode and common mode interference work together in the range of 0.1 to 1 MHz, and the common mode interference mainly works in the range of 1 to 30 MHz. Therefore, the magnetic properties of the filter inductors are completely different. For common mode inductors, materials with higher relative magnetic permeability should be selected, generally the relative magnetic permeability should reach about 15,000; while for differential mode, materials with lower relative magnetic permeability can be selected, generally about 10 to 100. Since the ground of the common mode capacitor is connected to the casing, for safety, the common mode capacitor cannot be too large, and a higher dielectric constant should be selected to enhance the voltage resistance of the capacitor. Table 1 lists the material parameters of different substances.

3 Analysis of equivalent parallel capacitance

Since the two windings of this LC planar magnetic integrated structure are very close, various noises in the power grid are often coupled into the circuit through the distributed capacitance between them. The best way to solve this problem is to add an electrostatic shielding layer between the primary and secondary windings as shown in Figure 11, where C1 and C2 are the distributed capacitances between the primary and secondary windings and the electrostatic shielding layer, respectively.

Considering the equivalent parallel parasitic capacitance of a real inductor, its equivalent impedance is:

If an item ω2L2C2 is added to the denominator of equation (1) (and ω2L2C2=ω2L1C1 is satisfied), then Z=jωL1, which will become an ideal inductor. Based on this idea, an inductor can be divided into two parts, and the midpoint is connected to the ground. Its model and equivalent circuit are shown in Figures 12 and 13 respectively. Decoupling and Y/△ transformation are performed on Figure 13, and the equivalent circuits obtained are shown in Figures 14 and 15 respectively. And its transformation parameters are:

If Gg=4Ce, Z12=1/Y12=jωL, the inductor will become an ideal inductor, the parasitic coupling capacitance between the windings will be reduced to zero, and our desired goal can be achieved. The additional capacitance Cg can be realized by adding an external capacitor or by using the parasitic capacitance between the winding and the ground.

For the planar LC magnetic integrated structure, in order to obtain the desired Gg, a ground layer can be added between the two windings, and its planar structure is shown in Figure 16. When designing, you can use: PlanaE43/10/28-3F and PLT43/28/4-3F3, the winding uses two layers, each layer has 3 turns. The winding width is 2mm, and the insulation layer thickness is 0.07 mm. Its equivalent circuit is shown in Figure 17. If the winding loss and core loss are ignored, L1, Cp1, Rp1t and Rs1 are the inductance and parasitic parameters of the first half of the inductor respectively; L2, Cp2, Rp2 and Rs2 are the parameters of the other half of the inductor respectively; L3 and Rs3 are the ground formation inductance and resistance. When the ground formation resistance is ignored, the simplified circuit is shown in Figure 18. And:

If Cg=4Ce can be satisfied, then the parasitic capacitance between the coil windings can be reduced to zero.

4 Simulation Verification

In order to verify whether inserting a conductive layer can improve the high-frequency performance of the filter, and to verify the relationship between Ce and Cg, the ideal embedded conductive layer length Xo can be found and X can be used as a variable. The simulation results are shown in Figure 19. Then, differential-mode and common-mode simulation circuits are established, and the capacitance values ​​Ce and Cg are changed according to Table 1. The differential-mode insertion loss simulation results are shown in Figure 20, and the common-mode insertion loss simulation results are shown in Figure 21. Figure 22 is the simulation circuit for the common-mode insertion loss.

According to the simulation results, as X increases, the resonant frequency increases, and when X=24.89 mm, the differential mode insertion loss shows an ideal state. At this time, Cg=4Ce.

5 Conclusion

The simulation results show that the filter embedded in the conductive layer can remove the influence of EPC and has good high-frequency performance. The insertion loss of the filter can reach -60 dB above 30 MHz and has a trend of further reduction.