Fully Differential Amplifier Provides High Voltage, Low Noise Signals for Precision Data Acquisition Signal Chain

summary

A fully differential amplifier (FDA) has differential inputs and differential outputs, and its output common mode is independently controlled by the direct current (DC) input voltage. It is mainly used in the front end of analog-to-digital conversion in data acquisition systems to condition the signal to a suitable level for the next stage (usually an analog-to-digital converter (ADC)). FDAs are generally designed in a single chip with a small power supply voltage, so the output dynamic range is limited. This article will introduce the design method of a high-voltage, low-noise FDA with an adjustable common-mode output. This article also fully analyzes FDA noise and its impact on the overall signal-to-noise ratio (SNR) of the signal chain of a high-performance data acquisition system.

introduction

High voltage FDAs are suitable for applications that require wide output dynamic range and AC performance similar to high performance FDAs. For example, testing and evaluating precision data acquisition signal chains with wide input ranges may require high voltage FDAs. The output voltage range of most current FDAs is generally limited due to the small supply voltage. FDAs are suitable for driving the input of high performance ADCs, which typically require a single supply. FDAs have excellent AC performance, with excellent SNR and total harmonic distortion (THD). However, FDAs are inferior to many higher voltage precision op amps in terms of offset, rail-to-rail swing, bias current, and drift performance. However, this is not a problem at all, as it meets the ADC driving requirements, and Analog Devices provides a range of ADC drivers for various applications.

FDAs accept single-ended or differential inputs, have gain, and provide differential outputs whose common mode can usually be adjusted via the output common-mode input pin (VOCM) (see Figure 1). The advantages of FDAs are greater output dynamic range, a maximum output of twice the output rail, and lower noise and even-order harmonic distortion. For example, the maximum output peak-to-peak of a ±5 V FDA is close to ±10 V or 20 V pp.

The output of the ±18 V circuit is greater than 60 V pp. The ADA4625-1/ADA4625-2 are low noise JFET amplifiers with very good noise and distortion performance and a wide supply range of ±18 V. Designing an FDA using a discrete op amp can be challenging when all the dc and ac performance requirements of the application need to be met.

Figure 1. FDA

To create a differential amplifier, a simple approach is to use a non-inverting and an inverting amplifier to produce a differential-mode signal at the output (Figure 2). However, the disadvantage of this approach is that the two amplifiers, U1 and U2, do not operate in a very symmetrical manner, so performance is not optimized.

Figure 2. Single-ended to differential conversion circuit

A better approach is to configure the two op amps differentially, similar to a basic differential amplifier, where U1 and U2 share feedback and gain resistors, with a gain of Av = (RG + 2RF)/RG (see Figure 3).

Figure 3. Differential amplifier circuit

This configuration provides balanced outputs with a simplified gain network, and the gain can be easily adjusted with the gain setting resistor RG. However, when the input is single-ended, the differential output will be asymmetrical in amplitude (see Figure 4). Asymmetrical outputs severely limit the output range because one output reaches the supply rail before the other. This problem can be solved by adjusting the resistor gain network to make the output symmetrical (Figure 5). Note that the gain resistor is split into two parts, RG1 and RG2, and U2 takes feedback from the center of RG1 and RG2, making the output symmetrical. The gain is given by: Av = (RG1 + RG2 + RF1 + RF2)/RG1.

Figure 4. Asymmetric output

Figure 5. Symmetrical output

Adding adjustable output common mode

There are two ways to add adjustable common-mode: one method is to use two ADA4625 devices to add a VOCM amplifier to each input (Figure 6 and Figure 7); the other method is to use only one ADA4625-1 as the VOCM amplifier (Figure 8 and Figure 9). Each of these methods has advantages and disadvantages, which are discussed in detail below.

By adding amplifiers U3 and U4, any DC input voltage (V6) applied is added to both the positive and negative inputs. Since the same voltage is added to each input, they appear DC common mode at the output. However, U3 and U4 will cause additional power dissipation in the circuit, in addition to the additional noise further amplified by the U1 and U2 differential stages. However, it is very simple and does not affect the overall signal gain. For the circuit in Figure 6, the signal gain is Av = (RG1 + RG2 + RF1 + RF2)/RG1; for the circuit in Figure 7, the signal gain is Av = (RG + RF1 + RF2)/RG.

Figure 6. Single-ended to differential adjustable common-mode circuit using dual amplifiers. The right figure shows the LTspice® simulation of the input (red) and output (blue and green).

Figure 7. Differential-to-differential adjustable common-mode circuit using dual amplifiers. The right image shows the LTspice simulation of the input (red) and output (blue and green).

Another way to add adjustable VOCM is to add an amplifier and add its output to each input. The advantages of this approach include fewer components (only one amplifier) ​​and resistors, as well as lower noise contributions from the added components. In fact, U3 does not generate any additional noise because its output-referred noise appears common-mode to the inputs of U1 and U2, in addition to the noise from the resistor divider R4 to R7.

Resistors R3 through R7 form a resistor adder network that adds VOCM to the input signal. R3 through R5 add the common mode to the positive input signal, while R6 through R8 (R6 and R7 for single-ended inputs) add the common mode to the negative input. Note that this resistor network attenuates the input signal. This reduces the overall signal gain of the circuit. The total signal gain is Av = [(RG1 + RG2 + RF1 + RF2)/RG1][(R4//R5)/(R4//R5 + R3)] for the circuit in Figure 8 and Av = [(RG + RF1 + RF2)/RG][(R4//R5)/(R4//R5 + R3)] for the circuit in Figure 9. The Noise Analysis section clarifies the major noise sources and discusses whether the second method of adding VOCM is more beneficial than the first method, depending on the desired overall gain and other factors that are important for the designer to consider.

Figure 8. Single-ended to differential adjustable common-mode circuit using a single amplifier. The right image shows the LTspice simulation of the input (red) and output (blue and green).

Figure 9. Differential-to-differential adjustable common-mode circuit using a single amplifier. The right image shows the LTspice simulation of the input (red) and output (blue and green).

Noise Analysis

Noise is a key consideration when providing stimulus for a high performance precision data acquisition signal chain and will ultimately determine the limits of the system in terms of dynamic range and SNR. The theoretical SNR for a 16-bit ADC is 98 dB (6.02 N + 1.76 dB, N = number of bits), which means that the equivalent noise for a 4.096 Vp output (or 8.192 V pp) is approximately 36 μV rms. This noise is called quantization noise and is caused by the quantization error of the ADC. -98 dB SNR is the ideal limit for a 16-bit system, and any degradation in performance will be caused by additional noise at the input of the ADC or in the surrounding circuitry. Below is an analysis of the noise contribution of each component in the fully differential circuit for single and dual amplifier VOCMs. Figure 10 shows the noise model for the FDA circuit with dual amplifier VOCMs.

Differential Stage—U1 and U2 Noise Contributions

The ADA4625-1/ADA4625-2 has a very low current noise density of 4.5 fA/√Hz at 1 kHz, while the voltage noise referred to the input (RTI) is approximately 3 nV/√Hz at 1 kHz, which is considered broadband noise for this analysis. The total noise contribution (rms) of the current and voltage noise of U1 and U2 at the differential output can be expressed as:

Where eNv,U1U2 is the output voltage noise due to the RTI voltage noise of U1 and U2, and eNI,U1U2 is the output voltage noise due to the input current noise. The RTI voltage noise is obtained by taking the square root of the sum of the squares (RSS) of the input noise components, which is then amplified by the gain and feedback networks RF and RG. Similarly, the current noise is RSSed and converted to voltage noise by RG, which is then amplified and transmitted to the output. The input current noise is very small and its contribution is negligible, so the voltage noise of the resistor and amplifier is the main noise source at the output.

Figure 10. Dual-amplifier VOCM noise model

The output noise due to the gain of U1 and U2 and the feedback resistor network (RF1, RF2, and RG) is:

Here, the thermal noise of a 1 kΩ resistor at room temperature is 4.06 nV⁄√Hz.

Combining the voltage noise of U1 and U2 and the noise of their feedback resistor network at the output, and neglecting the current noise, we can use Equations 1 and 3 to obtain:

As we have seen from the previous discussion, as the gain is increased, the voltage noise of the amplifier can easily become the dominant noise. Using a smaller value of RG (such as 500 Ω) can greatly reduce the noise of the resistor.

VOCM Circuit—U3 and U4 Noise

Next, we analyze the noise of the VOCM circuit in Figure 10. The total noise of the VOCM circuit (U3 and U4), including the resistor noise and ignoring the input current noise of each amplifier, is calculated as follows:

Total VOCM output noise =