Research on bipolar phase-shift controlled high-frequency pulse AC link inverter

The high-frequency pulse AC link inverter circuit topology family and its bipolar phase-shift control strategy are proposed and studied in depth. With the help of cycloconverter commutation overlap and output filter inductor current polarity selection, the bipolar phase-shift control strategy realizes the natural commutation of transformer leakage inductance energy and filter inductor current, solves the inherent voltage overshoot of this type of inverter and the circulation phenomenon of cycloconverter during commutation overlap, and realizes zero voltage switching of inverter bridge power devices and zero current switching of cycloconverter power devices. Both simulation and principle test results confirm the feasibility of this bipolar phase-shift control strategy and the correctness of theoretical analysis.

Keywords: high-frequency pulse AC link; bipolar phase-shift control; zero voltage and zero current switch; cycloconverter; commutation overlap

0 Introduction

Although traditional inverter technology is mature, reliable and widely used, it has disadvantages such as large and heavy size, high audio noise and poor system dynamic characteristics [1]. Replacing the industrial frequency transformer in the traditional inverter with a high-frequency transformer overcomes the disadvantages of the traditional inverter and significantly improves the characteristics of the inverter. The high-frequency pulse AC link inverter [1][2] has the characteristics of bidirectional power flow, two-stage power conversion (DC/HFAC/LFAC), high conversion efficiency and reliability, but has disadvantages such as voltage overshoot during commutation of the cycloconverter device. It is usually necessary to use a buffer circuit or an active voltage clamp circuit to absorb the energy stored in the leakage inductance, thereby reducing the conversion efficiency or increasing the complexity of the circuit.

Therefore, how to solve the inherent voltage overshoot phenomenon of high-frequency pulse AC link inverter and realize soft commutation of cycloconverter without increasing the complexity of circuit topology is the research focus of this type of inverter.

1 High-frequency pulse AC link inverter circuit topology family

The high-frequency pulse AC link inverter circuit topology family is shown in Figure 1. This type of circuit consists of a high-frequency inverter, a high-frequency transformer, and a cycloconverter, and has the advantages of simple circuit topology, two-stage power conversion (DC/HFAC/LFAC), bidirectional power flow, and high conversion efficiency.

The push-pull circuits shown in Figures 1(a) and 1(b) are suitable for low-voltage input conversion applications, and the bridge circuits shown in Figures 1(c) to 1(f) are suitable for high-voltage input conversion applications; the full-wave circuits shown in Figures 1(a), 1(c) and 1(e) are suitable for low-voltage and high-current output applications, while the bridge circuits shown in Figures 1(b), 1(d) and 1(f) are suitable for high-voltage and low-current output applications.

(a) Push-pull full-wave

(b) Push-pull bridge

(c) Half-bridge full-wave

(d) Half-bridge

(e) Full-bridge full-wave

(f) Full bridge type

Figure 1 High-frequency pulse AC link inverter circuit topology family

2 Steady-state analysis of bipolar phase-shift controlled high-frequency pulse AC link inverter

2.1 Bipolar Phase Shift Control Principle

Taking the full-bridge full-wave circuit topology as an example, its bipolar phase-shift control principle is shown in Figure 2. The output voltage uo is compared with the sinusoidal reference voltage uref, and the error amplification signal ue is obtained through the PI regulator. After ue is compared with the two carrier signals uc1 and uc2 with opposite polarities, it is divided by two on the rising edge, and then turned on according to the output filter current polarity to obtain the drive signals of switches S5 and S6. The drive signals of switches S7 and S8 are respectively complementary to the signals of S5 and S6, and there is a commutation overlap time (not shown in Figure 2). The drive signals of switches S1 and S4 are obtained by dividing the carrier signal uc1 by two, and the drive signals of switches S2 and S3 are obtained after inversion.

(a) Circuit topology

(b) Bipolar phase shift control principle

Figure 2 High-frequency pulse AC link inverter circuit topology and its bipolar phase shift control principle

The power switches S5 and S7 (S6 and S8) of the cycloconverter have a commutation overlap conduction time, and the power switches S5 and S6 (S7 and S8) are selectively turned on according to the polarity of the filter inductor current iLf, so that the control scheme has the following advantages:

1) During the commutation overlap period of the cycloconverter, the natural commutation of the transformer leakage inductance energy is realized, the zero current switching of the power device is realized, and the inherent voltage overshoot phenomenon is solved;

2) The natural continuous flow of the filter inductor current is realized;

3) The introduction of the filter inductor current polarity selection signal avoids the circulation phenomenon in the cycloconverter during the commutation overlap period;

4) Energy feedback from the AC side twice in each switching cycle enables zero-voltage switching of all power devices in the inverter bridge.

The driving signals between power switches S5, S6 and S1, S4 (S7, S8 and S2, S3) all have a phase difference θ (0≤θ≤180°), and the common conduction time DTs/2 in one switching cycle can be expressed as

DTs/2=Ts(180°-θ)/(2×180°)(1)

Where: Ts is the switching period.

Since the phase shift angle θ and the common conduction time DTs/2 both change according to the sinusoidal law, and the output filter front-end voltage uDC is a bipolar SPWM wave, this control method is called bipolar phase shift control. Adjusting the phase shift angle θ can achieve the stability of the output voltage when the input voltage or load changes.

2.2 Steady-state analysis

Assume that the leakage inductance of the primary and secondary sides of the transformer are equal, that is, Llk1 = Llk2 = Llk3 = Llk. The inverter has 12 operating modes in one switching cycle, as shown in Figure 3.

(a) Steady-state principle waveform within a switching cycle

(b) t = t1 to t2

(c) t = t2 to t3

(d) t = t3～t4～t5

(e) t = t5 to t6

(f) t = t6～t7～t8

Figure 3 Steady-state principle waveform within a switching cycle

1) t=t1～t2: At t1, the power switches S1 and S4 realize ZVS opening, the output filter inductor current iLf continues to flow through the power switches S7 and S8, and the AC side energy is fed back to the DC power supply through D1 and D4, as shown in Figure 3 (b).

2) t = t2 ~ t3: At t2, S5 realizes ZCS opening. During this commutation overlap period, iLf flows through S7, S8 and S5, S6, i2 increases rapidly, i3 decreases rapidly; i1 quickly changes from negative to positive, as shown in Figure 3 (c). Assuming the induced electromotive force of the primary winding of the transformer is e, then

e=Ui－Llk1 =uACN1/N2=－uBCN1/N2 (2)

uAC－Llk2 =uDC=Lf +uo（3）

uBC－Llk3 =uDC=Lf +uo（4）

i2＋i3=iLf（5）

i1=(i2－i3)N2/N1＋iM（6）

Assuming that the magnetizing inductance LM and the output filter inductance Lf are much larger than the leakage inductance, the magnetizing current iM is negligible, and the change rate of iLf is very small during the commutation overlap period, we can get

-uAC+2Llk -Llk +uBC=-2 e+2Llk =0 (7)

e=Ui－2 Llk + Llk -Llk =Ui－2 Llk (8)

e= = Llk =－ Llk (9)

From formula (9), we can see that the change rate of i2 and i3 is N1Ui/(3N2Llk), the change rate of i1 is 2Ui/(3Llk), and the potential of points D and C is equal. When i2 rises to the iLf value, i3 drops to zero. Due to the blocking of switch S8, i3 cannot increase negatively after dropping to zero, and formula (9) is no longer valid. ZCS soft commutation is achieved between switches S7 and S5. From formula (9), we can see that the commutation overlap time tco is

tco(>=)t3－t2=3ILfm （10）

Where: ILfm is the peak value of the filter inductor current at rated load.

3) t=t3～t4: At t3, the soft commutation between switches S5 and S7 ends. iLf flows through S5 and S6, i1 flows through S1 and S4, and energy is transferred from the DC side to the AC side, as shown in Figure 3(d).

4) t=t4～t5: At time t4, switch S7 is turned off with zero current, as shown in Figure 3(d).

5) t=t5～t6: At t5, switches S1 and S4 are turned off by ZVS, C1 and C4 are charged, and C2 and C3 are discharged. The drain-source voltages uDS2 and uDS3 of switches S2 and S3 decrease, as shown in Figure 3(e).

6) t=t6～t7: At t6, uDS2 and uDS3 drop to zero, then i1 continues to flow through D2 and D3, and the transformer primary leakage inductance energy and AC side energy are fed back to the DC power supply, as shown in Figure 3 (f). At t7, S2 and S3 are turned on at zero voltage.

The working process of half a switching cycle after time t7 is similar to the first half and its switching state is equivalent to the circuit of the switching cycle.

3 Simulation and principle test

Design example: full-bridge full-wave circuit topology, bipolar phase-shift control strategy, rated capacity S = 1kVA, input voltage (DC) Ui = 270 (1 ± 10%) V, output voltage (AC) Uo = 115V, output voltage frequency fo = 400Hz, load power factor -0.75 ~ 0.75, switching frequency fs = 50kHz, turns ratio N1/N2 = 22/22, filter inductor Lf = 1mH, filter capacitor Cf = 4.7μF/250V. 3.1 Simulation results and discussion

The steady-state simulation waveforms at different input voltages and different loads are shown in Figure 4. In Figure 4 (e), uGS1, uGS2, uGE5, and uGE7 are the drive signals of power switches S1, S2, S5, and S7, respectively. The filter front-end voltage uDC is a three-level bipolar SPWM wave; power switches S1 to S4 achieve ZVS, and power switches S5 to S8 achieve ZCS; the inverter has good load adaptability and voltage regulation performance. The simulation results are consistent with the theoretical analysis.

(a) Rated input voltage, rated resistive load

(b) Rated input voltage, no-load

(c) 90% rated input voltage, rated inductive load

(d) 110% rated input voltage, rated capacitive load

(e) ZVZCS switching waveform

Figure 4 1kVA bipolar phase-shift control inverter simulation waveform

3.2 Experimental results and discussion

1kVA DC 270V/AC 115V 400Hz bipolar phase-shift control high-frequency pulse AC link inverter consists of three parts: power circuit, control circuit, and internal auxiliary power supply. The control circuit is mainly composed of reference sine wave circuit, error amplifier circuit, inductor current polarity judgment circuit, control signal generation circuit (2 UC3879 phase-shift control chips) and drive circuit. Switches S1~S4 use IRFP460 MOSFET (20A/500V), switches S5~S8 use HGTG10N120BND IGBT (35A/1200V), and the drive circuit uses A3120 chip.

The principle test waveform is shown in Figure 5. Near the zero-crossing point of the output filter inductor current, the output voltage waveform is distorted, which is caused by the introduction of the current polarity selection signal by the cycloconverter. The test results confirm the feasibility of this type of inverter.

Vertical axis: uO 100V/div, iLf 13.5A/div;

Horizontal axis: t 400μs/div

Figure 5 Principle test waveform

4 Conclusion

1) High-frequency pulse AC link inverter topology family, including 6 circuits such as push-pull full-wave type.

2) With the help of cycloconverter commutation overlap and output filter inductor current polarity selection, the bipolar phase-shift control strategy realizes the natural commutation of transformer leakage inductance energy and filter inductor current, solves the inherent voltage overshoot and circulating current phenomena, and realizes inverter bridge ZVS switching and cycloconverter ZCS switching.

3) Both simulation and principle test results confirm the feasibility of this phase-shift control strategy and the correctness of the theoretical analysis.