How to use a high precision instrumentation amplifier for remote sensing?

When it comes to sensors, few things beat the Wheatstone bridge (Figure 1). The bridge produces a differential voltage that changes predictably as a physical parameter changes. The differential voltage also has the added benefit of rejecting drift over temperature and time. The differential voltage sits on top of a larger common-mode (CM) voltage. An instrumentation amplifier is used to amplify the small signal provided by the bridge. The benefit of an instrumentation amplifier is that it can sense the differential voltage and reject CM to a degree that is impossible with a traditional op amp (due to the requirement for highly matched external resistors) when there is little or no loading on the bridge elements.

Figure 1. Wheatstone bridge

The electronics used to make physical measurements are often located far from the physical parameter being measured. For example, strain gauge measurements buried under the pavement at a truck weigh station or within a bridge structure are unlikely to be located next to the electronics that read the measurements. When using a two-wire quarter-bridge strain gauge (such as Omega's SGT-1/350-TY43), the sensor is placed far away from the sense amplifier, as shown in Figure 2, which produces unsatisfactory results, even when shielded twisted-pair cable is used for the sensor leads.

Figure 2. Remote sensor setup is affected by ambient noise pickup.

The problem is that shielded twisted pair is not immune to all interference on long cable runs. In this case, the instrument’s well-balanced inputs cannot be relied upon to eliminate CM effects. The interference picked up by the long cable affects the positive and negative amplifier inputs unevenly, and the inputs contain uncorrelated signals that CMR cannot eliminate. Therefore, it is not surprising to find significant noise at the circuit output, as shown in Figure 3, due to the unbalanced response to (what appears to be) CM noise.

Figure 3. Troublesome 120 Hz noise at the amplifier output (0.1 V/div, 2 ms/div)

To successfully extract the small bridge differential voltage from CM (DC and interference), one solution is to use two pairs of shielded or unshielded twisted pair (UTP). In this way, the two inputs of the instrumentation amplifier are balanced and equally affected by CM noise, as shown in Figure 4. Devices such as the LT6370 have excellent low-frequency CMR (120 dB) and can reliably reject the noise that plagues the IA input. As a result, the output waveform is clean over long distances even in noisy environments.

Figure 4. Remote sensing using two unshielded twisted pairs.

With the full CMR capabilities of the LT6370, we can go a step further and simplify the configuration by reducing one pair of wires, leaving only one UTP. This concept is shown in Figure 5, where the input to U2 is balanced for good CMR. Note that the UTP leads look identical to U2 and have the same impedance to ground (R2, R4).

Figure 5. Single UTP root for remote sensing

For the component values ​​shown in Figure 5, the current flowing through the sensor, R _SENSOR, is approximately 1 mA. Using the RG1 value for U 1, the stage operates with G = 10 V/V, and the output voltage is a 10x amplified copy of the voltage across RSENSOR, approximately 3.5 V. The primary task of U1 is to remove interference that exists on the long UTP wire and responds only to the sensor voltage, which is equal to the sensor resistance multiplied by the approximately 1 mA current flowing through it. The LT6370’s exceptionally low offset voltage and drift, along with its excellent CMR characteristics, made it an obvious choice.

The other half of the Wheatstone bridge consists of R5, R6, and VR1, which has almost the same current as the sensor portion of the bridge. The sensor voltage at the output of U1 and the reference voltage at the VR1 wiper are both low-pass filtered to the differential inputs of U2 to eliminate interfering noise. U2 is set to high gain (G = 1 + 24.2 kΩ/RG2 = 100 V/V) to amplify the very small sensor voltage on the positive input, while the negative input is a fixed low-noise reference voltage generated from the reference voltage source LT6657-5. The output of U1 accurately represents the measured strain applied to the sensor (which is attached to the target component or material) to drive an ADC or other similar signal processing.

An optional DAC and OPA (U4, U5) connected to the REF pin of U2 (which can be grounded if no offset adjustment is required) can be used to provide output offset adjustment and zeroing. The U2 output voltage can be shifted to a reference or CM level appropriate for the selected ADC using the DAC. For example, an ADC with a 5V reference can be driven directly from U2, using the DAC to drive the U2REF input to set its zero output to 2.5V. In this way, a 0V to 2.5V ADC analog input represents compressive strain, and a 2.5V to 5V signal represents tensile strain. It is important to note that the device driving the U2 REF pin (in this case, the AD820) should maintain low impedance to eliminate any possible gain errors.

The following are expressions for the relationship between the output voltage and the sensor resistance and the relationship between the output voltage and the measured strain (ε):

Where ΔR _SENSOR is the change in sensor resistance caused by strain

in:

L refers to the sensor length

ε refers to the measured strain

For the selected sensor:

R _sensor = 350 Ω

GF = 2

The resulting strain (ε) is:

The LT6370's ultra-low gain error (less than 0.084% at G = 10 V/V) and low input offset voltage (less than 50 μV maximum over temperature) ensure that U2 obtains a true copy of the sensor voltage, minus the interference picked up by the UTP, and compares it to the reference voltage generated at the inverting input of U2. The LT6657-5 generates a stable, low noise, low drift reference voltage, making the entire circuit immune to power supply voltage variations. Of particular importance, the LT6657-5's 1/f noise is very low, which is significant because the circuit has a large gain.

The simple RC low-pass filters (R9, C2 and R10, C3) on each input of U2 are set with a roll-off frequency of approximately 10 Hz, and the output noise can be reduced by limiting the bandwidth.

As shown in Figure 6, the LT6370 has a low 1/f noise corner frequency (<10 Hz), and the effect of 1/f noise is small, which is an advantage. In addition, the current noise density plot shows that it is much better to keep the two input impedances balanced to minimize the current noise effect, using the correlated components of the noise at the input. Therefore, since the wiper of VR1 has an equivalent impedance, the value of R10 is reduced to 3.74 kΩ to match the 4.75 kΩ impedance of R9.

Figure 6. LT6370 input referred current/voltage noise density.

The bridge sensor is located far from the signal processing amplifier, requiring an instrumentation amplifier to extract the clean measured differential voltage. The characteristics of the LT6370 instrumentation amplifier enable it to successfully process signals from remote sensors over long cables. The LT6370 manufacturing process uses an on-chip heater during production test to guarantee temperature drift values, further enhancing the LT6370's suitability for remote monitoring applications and extending its service life and product life in hard-to-repair equipment.