Multi-resonant converter with a switching frequency of 1MHz

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Multi-resonant converter with a switching frequency of 1MHz [Copy link]

A multi-resonant DC/DC converter that is very suitable for working at ultra-high frequency is analyzed. All the switch tubes of the converter work in the ZVS state, and all the rectifier diodes work in the ZCS state. The converter has a simple structure and only needs one magnetic component for the entire converter. The ultra-high frequency adaptability of the converter is analyzed in detail. A prototype with 135V input, 54V/3A output, and a switching frequency higher than 1MHz verifies its working principle and ultra-high frequency adaptability. The prototype has an efficiency of 88.7% under rated conditions.

Keywords: multi-resonance; soft switching; converter

　

0 Introduction

Miniaturization is the goal of current power products. Increasing the switching frequency can reduce the size of components such as inductors and capacitors. However, the bottleneck of increasing the switching frequency is the switching loss of the switching devices, so soft switching technology came into being. At present, with the application of soft switching technology, the switching frequency of the converter can easily exceed 100kHz. Soft switching circuits can be divided into two types: buffer type and control type. The buffer type soft switching topology often adds a lot of extra circuits, which increases the cost and reduces reliability, making it difficult for users to accept. The control type soft switch does not increase the components of the main circuit. It can achieve soft switching through a reasonable design of the control circuit, which is easier for users to adopt. At present, there are not many mature control type soft switching circuits, and typical ones include phase-shifted full bridge [1] and asymmetric half bridge [2]. These are PWM type converters that achieve soft switching through edge resonance, which can reduce switching losses without increasing the effective value of voltage or current. However, it is difficult for this type of circuit to truly achieve soft switching of all semiconductor devices (including switches and diodes). For example, the rectifier diodes of the phase-shifted full bridge and asymmetric half bridge are hard-off, which has a serious reverse recovery problem. Therefore, these circuits cannot operate at higher switching frequencies. Therefore, when the switching frequency needs to be further increased, it is more suitable to use a resonant converter.

A multi-resonant DC/DC converter is proposed below, with a switching frequency exceeding 1MHz. All semiconductor devices of this converter realize soft switching, which is a good choice for ultra-high frequency converters.

1 Working Principle

FIG1 shows an LLC series multi-resonant converter with a half-bridge structure: two main switches S1 _and S2 _form a half-bridge structure, and the driving signal is a complementary signal with a fixed duty cycle of 50%. The constant output voltage is achieved by changing the switching frequency. Therefore, this type of resonant converter can also be classified as a controlled soft switching circuit. The inductor _Ls , the capacitor Cs and _the excitation inductor Lm of the transformer _form an LLC resonant network. The resonant network is connected between the midpoint of the half-bridge and the ground, so the resonant capacitor Cs also acts as a DC blocking capacitor. On the output side, the rectifier diodes D1 _{and D2 form} a _center -tapped rectifier circuit, and the rectifier diodes are directly connected to the output capacitor Co.

Figure 1 LLC series multi-resonant DC/DC converter

The natural resonant frequency of LC is defined as

fs =1/(2π ₎（1）

The natural resonant frequency of LLC is defined as

f _m =1/(2π )（2）

The switching frequency range of the LLC series multi-resonant converter described in this article is f _m < f < f _s .

In the following analysis, C _o is considered to be infinite and replaced by a constant voltage source V _o , and the main switch has a reverse parallel diode. One switching cycle of the converter can be divided into 6 working stages, and its equivalent circuit is shown in Figure 2. The corresponding working waveform is shown in Figure 3. The working principles of the 6 working stages are as follows.

(a) Stage 1 [ t ₀ , t ₁ ]

(b) Phase 2 [ t ₁ , t ₂ ]

(c) Stage 3 [ t ₂ , t ₃ ]

(d) Stage 4 [ t ₃ , t ₄ ]

(e) Stage 5 [ t ₄ , t ₅ ]

_{( f )} Stage 6 [ t5 , t6 ]

Figure 2 Equivalent circuits at each stage

Figure 3 Main working waveform

1) Phase 1〔t ₀ ~ t ₁〕 At t ₀ , S ₂ is turned off, the resonant current i _r discharges the output capacitor of S ₁ , and the drain-source voltage v _ds1 of S ₁ begins to decrease. When v _ds1 drops to zero, the body diode of S _{1 is turned on. The input voltage is applied to} the LLC series loop. On the secondary side, the polarity of the transformer winding is positive at the top and negative at the bottom, D ₁ is turned on, and the voltage of L _m is clamped by the output voltage V _o . The resonance actually occurs between L _s and C _s , and the _current im on L _m increases linearly.

_{2 )} Phase 2〔t1～t2〕 At t1 _, S1 is turned on under zero voltage condition. i _m continues to rise linearly, i _r flows through S1 _and gradually rises in the form of a sine wave. The output current flowing through D1 _is the difference between the resonant current and the excitation current. The switching period is greater than the resonant period of L _s and C _s , so after i _r passes half the resonant period, S1 _is still in the on state. When i _r drops to be equal to i _m , the D1 _current is turned off due to zero crossing. This working stage ends.

Since the voltage applied to L _{m is} nV _o , i _m can be expressed as

i _m ( t ) = t－I _m（3）

Where I _m = (4)

Where: I _m is the maximum value of the excitation current;

V _o is the output voltage;

n is the turns ratio of the primary to the secondary side of the transformer.

3) Phase 3〔t ₂ ~ t ₃〕 At t ₂ , D ₁ is turned off under zero current condition. The output side is completely separated from the resonant circuit. The voltage of L _{m is no longer limited by} V _o , and L _m and L _s are connected in series to participate in the resonance. In the usual circuit design, L _m >> L _s , so the resonant period becomes significantly longer. i _r remains basically unchanged, and it can be considered that

i _r ( t ) = i _m ( t ) = I _m (5)

During this stage, i _r continues to charge C _{s , and the voltage of} C _s continues to rise until moment t ₃ , when S ₁ is turned off and the next half working cycle begins.

Working stages 4, 5, and 6 are similar to working stages 1, 2, and 3. The difference is that the initial energy of resonance is provided by the resonant capacitor _Cs . The working waveform is completely symmetrical with stages 1, 2, and 3.

4) Phase 4〔t ₃ ~ t ₄〕 At t ₃ , S ₁ is turned off, i _r discharges the output capacitance of S ₂ , and the drain-source voltage v _{ds2 of S 2} begins to decrease. When v _ds2 drops to zero, the body diode of S ₂ is turned on. On the secondary side, the polarity of the transformer winding is negative at the top and positive at the bottom, D ₂ is turned on, the voltage of L _{m is clamped by} V _o , and the resonance actually occurs between L s _and C s _{. The current} im on L _m decreases linearly.

_{5 )} Stage 5〔t4～t5 _〕 At t4 _, S2 is _turned on under zero voltage condition. i _m continues to decrease linearly, i _r flows through S2 _and increases negatively in the form of a sine wave. The output current flowing through D2 _is the difference between the resonant current and the excitation current. In this operating frequency range, the switching period is greater than the resonant period of L _s and C _s . Therefore, after i _r has gone through half a cycle of resonance, S2 _is still in the on state. When i _r drops to be equal to i _m , D2 _current crosses zero and is turned off. This working stage ends.

_{6 )} Phase 6〔t5～t6 _〕 At t5 _, D2 is turned off under zero current condition. The output side is completely separated from the resonant circuit. _The voltage of Lm is no longer limited by V0 , and _Lm and _{Ls are} connected in series to participate in the resonance. i _r remains basically unchanged and continues to discharge the resonant capacitor Cs _{. The voltage of} Cs _continues to decrease until t6 , when S2 _{is turned} off and a new working cycle begins.

Assuming that i _r remains constant from t ₂ to t ₃ and from t ₅ to t ₆ and is represented by _Im , the output voltage _Vo can be expressed as

V _o = V _in＋( T T _s ) (6)

Where: Vin _is the input voltage;

T is the switching period;

Ts _{is the} resonance period of Cs and Ls , Ts = 1 / _{fs =} 2π _.

It can be seen from equation (6) that the output voltage increases with the increase of the switching period.

2 High frequency adaptability analysis

The LLC multi-resonant converter analyzed above is very suitable for applications with very high switching frequencies for the following reasons.

1) All switches work in the ZVS state, and the switching loss is almost zero. The zero voltage of the switch is achieved by the magnetizing current on the magnetizing inductor charging and discharging the junction capacitance of the switch. Therefore, for changes in load current, the zero voltage turn-on condition will basically not change, which is better than other control-type soft PWM circuits such as phase-shifted full-bridge. In addition, the magnetizing inductor of the LLC multi-resonant converter is used as one of the resonant inductors to adjust the relationship between the input and output voltages, and it is designed to be relatively small. From the perspective of conduction loss, this is disadvantageous, but from the perspective of the realization conditions of soft switching, it is very advantageous. Therefore, this circuit is very advantageous in ultra-high frequency applications. The limit condition of ZVS is shown in formula (7) (the limit condition means that the critical condition of ZVS can be achieved by assuming that the dead time can be arbitrarily large).

(7)

Where: C _oss1 and C _oss2 are the output capacitances of the two switching tubes respectively.

Substituting equation (4) into equation (7), we can obtain the further expression of the limit condition of ZVS as equation (8).

(8)

In fact, in the LLC multi-resonant converter, equation (8) is very easy to satisfy, and the dead time will not be very large. Therefore, it can be approximately considered that the current on the magnetizing inductor remains unchanged during the dead time, that is, a constant current source charges and discharges the junction capacitance of the switch tube. The ZVS condition in this case is called the margin condition, and the expression is equation (9).

I _mtdead (>=)（C _oss1 + C _oss2 ) V _in (9)

Where: t _dead is the dead time.

Substituting equation (4) into equation (9), we can obtain the further expression of the ZVS margin condition as equation (10).

(10)

2) All secondary diodes work in ZCS state, and the impact of reverse recovery is small. However, ordinary control-type soft PWM circuits only realize soft switching of the switch tube, but do not solve the reverse recovery problem of the diode well. Therefore, it is still difficult to use in situations with very high switching frequencies (such as above 1MHz). The current waveform of the secondary diode is approximately sinusoidal, which is a disadvantage for reducing conduction loss. However, in ultra-high frequency applications, switching loss is much more difficult to handle than conduction loss. Therefore, this circuit has an advantage in ultra-high frequency applications.

3) Another reason why ordinary control soft PWM circuits are difficult to operate above 1MHz is that transformer leakage inductance is difficult to handle at high frequencies. Especially when considering the insulation strength of the primary and secondary sides, the transformer leakage inductance is difficult to reduce, and at ultra-high frequencies, the impact of leakage inductance is very obvious. The leakage inductance of the LLC multi-resonant converter is one of the resonant inductors or a part of the resonant inductor, and it is hoped that the leakage inductance can be designed to be larger. In low-frequency situations, it is usually difficult to design the required leakage inductance and an external resonant inductor must be added, while in high-frequency situations, it is easier to design the required leakage inductance. Therefore, this is another reason why this circuit is suitable for ultra-high frequency situations.

3 Experimental Results

A DC/DC converter with a switching frequency above 1MHz verifies the working principle and high-frequency adaptability of the multi-resonant converter.

The specifications and main parameters of the converter are as follows:

Input voltage V _in 135V;

Output voltage V _o 54V;

Output current _Io 0 ～3A;

Minimum operating frequency f 1MHz;

Main switches S1 _and S2 _IRFP250 ;

Rectifier diodes D1 _and D2 _30CPQ150 ;

Transformer Tn = 13: (7 + 7), Lm = 15μH, Ls ₌ 6μH _;

The resonant capacitor Cs _is 4.4nF (the actual capacity of Cs is smaller than this value at high frequencies).

Figure 4 shows the conversion efficiency of the converter under different loads. Its highest efficiency reaches 89.5% and the full load efficiency reaches 88.7%.

Figure 4 Efficiency curves at different load currents

Figure 5 shows the main experimental waveforms when the input voltage is 135V. Figure 5 (a) shows the v _ds and v _gs waveforms of S ₂ at full load (3A) . It can be seen that the driving voltage v _{gs of S 2} starts to rise after the v _ds voltage drops to zero, so it is turned on at zero voltage. The v _ds and v _gs waveforms of S ₁ are similar and are not given here one by one. Figure 5 (b) shows the resonant voltage and current waveforms of the primary side. There are two resonant processes in each half cycle, namely the resonance of C _s and L _{s , and the resonance of} C _s and ( L _s + L _m ). Figure 5 (c) shows the voltage and current waveforms on the rectifier diode D _1. It can be seen that the current resonates to zero in a sinusoidal shape, but a certain reverse recovery current still appears. This is because the switching frequency is 1MHz. Although it is a sinusoidal current waveform, its di / dt is still quite large. If a general PWM circuit is used at the same frequency, the reverse recovery problem will be more serious. Therefore, using ordinary Schottky or fast recovery diodes, general PWM circuits cannot operate at a frequency of 1MHz. The diode voltage here will also overshoot due to reverse recovery, but its overshoot voltage still does not exceed 2 times the output voltage. Therefore, a 150V Schottky diode can be used here, which is impossible in general PWM circuits.

(a) v _ds and v _gs of S ₂

(b) Resonant voltage and current

(c) Voltage and current of D ₁

Figure 5 Experimental waveform ( f = 1MHz I _o = 3A)

Figure 6 shows the converter's operating waveform at low frequency (120kHz) for comparison. It can be seen that except for a certain reverse recovery current in diode D1 _, the operating conditions at 1MHz switching frequency are basically the same as those at 120kHz switching frequency, which shows that the circuit has very good ultra-high frequency adaptability.

(a) v _gs and v _ds of S ₂

(b) Resonant voltage and current

(c) Voltage and current of D ₁

Figure 6 Experimental waveform ( f = 120kHz I _o = 10A)