Research on Current Mode Controlled Current Doubler Rectifier ZVS PWM Full-Bridge DC-DC Converter-EEWORLD

Collect

1. Introduction
The traditional PWM DC/DC phase-shifted full-bridge zero voltage soft switching (ZVS) converter uses the resonance between the leakage inductance or/and the primary series inductance of the transformer and the external or/and parasitic capacitance of the switch tube to achieve zero voltage soft switching. Since the conditions for achieving zero voltage soft switching ZVS in the leading bridge arm and the lagging bridge arm are different, it is much more difficult for the lagging bridge arm to achieve zero voltage soft switching ZVS than the leading bridge arm; the reverse recovery of the diode turned off during the output rectifier diode commutation will cause a large voltage spike on the secondary side; and there is also a more serious loss of the secondary side duty cycle. In order to solve these problems, an improved circuit topology is proposed below.

2. Improved phase-shifted full-bridge ZVS DC-DC converter main circuit
The improved phase-shifted full-bridge ZVS DC-DC converter main circuit structure and waveform comparison at each point are shown in Figure 2-1 (a) and (b):

It is easy to see that the main difference between the improved circuit topology and the basic circuit lies in the secondary rectifier circuit, which is called the current-doubler rectifier (CDR), and is one of the hot spots in current applications. The following is an introduction to the rectifier circuit. Compared with full-wave rectification, the secondary winding of the high-frequency transformer of the current-doubler rectifier only requires a single winding, without a center tap. Compared with the bridge rectifier, the number of diodes used in the current-doubler rectifier is half. Therefore, the current-doubler rectifier is a new type of rectifier that combines the advantages of both full-wave rectification and bridge rectification. Of course, the current-doubler rectifier uses an additional small output filter inductor. However, the operating frequency and transmission current of this inductor are both half that of the full-wave rectifier, so it can be made smaller. In addition, the dual inductor is more suitable for the requirements of distributed power dissipation.

Next, let's study the differences in the working conditions of the converter main circuit after changing the rectifier circuit.

Since the circuit working state can be divided into two completely identical processes within one cycle, the following only analyzes the situation of half a cycle, and this half cycle can be divided into the following three switching modes (compare with Figure 2-1 above).

(1). Switching mode 1: t0 _< t < t1 _where t1 = DTs _/ 2

At this time, Q1 _and Q4 _are turned on at the same time, the transformer secondary inductor L1 _and rectifier tube _DS2 are turned on, and the primary energy is transferred to the load end. The equivalent circuit of this mode is shown in Figure 2-2:

Where a is the transformer ratio, Vin is the DC bus voltage, I1 _and I2 _are the inductor L1 and L2 currents respectively ₍ L1 ₌ L2 ₌ Ls ₎ , _and the following equation holds true:

This mode ends when Q4 turns off _.

(2). Switching mode 2: t ₁ < t < t ₂ where t ₂ ≤T _s /2

At time t1 _,_Q4 is turned off . At this time, the energy stored in the secondary inductor L1 _{charges
the capacitor}_Q4 (or parallel capacitor) and discharges the charge of the capacitors across Q3 _. In order to achieve soft switching, there must be at least a dead time Δt1 between _the turn-off of Q4 and the turn-on of Q3 _,_so that D3 is turned on first before Q3 _is turned on, and there is an _equation :

Established. Where C _eff is the equivalent capacitance across the drain and source of the switch tube, and I _p1 is the current flowing through the primary side of the transformer at time t _{1.
When D}_{3
is turned on, the two diodes}_DS1 and _DS2 on the secondary side of the transformer are turned on at the same time, and the circuit works in the freewheeling state. At this time, the equivalent circuit is shown in Figure 2-3 below:

At this time, the following circuit equation is established:

Where D is the pulse duty cycle, fS _is the circuit operating frequency, L' _ik is the primary transformer leakage inductance (or the series value with the external inductor), rt _is the transformer primary equivalent resistance, τ is the primary equivalent current decay time constant, and Vfp _is the anti-parallel diode conduction voltage drop.

(3). Switching mode 3: t ₂ < t < t ₃ where t ₃ =T _s /2

In this mode, the conduction condition of the primary side of the circuit is consistent with the above mode 2. At this time, since the commutation process is completed, DS2 _is turned off. Therefore, the equivalent circuit is shown in Figure 2-4 below:

At this time, the circuit equation is as follows:

Note that I1 _and I2 are the same as _in Mode 2, but the entire load current will flow through _DS1 . This mode ends when Q1 turns off. The energy stored in the secondary inductor L2 charges _and discharges the drain-source capacitance of the switches Q1 _and_Q2 at _the same time .

After Q ₁ is turned off, D ₂ and D ₃ will be turned on. At this time, Q ₂ and Q ₃ can be given an opening trigger signal. When the current reverses, Q ₂ and Q ₃ are turned on, and energy is transferred from the primary side to the secondary side again, so Q ₂ and Q ₃ are both turned on with zero voltage.

Due to symmetry, the remaining half cycle operates exactly the same as above.

From this we can get the output voltage at the load end:

Note that it is 1/2 times the general full-wave rectifier circuit.

The following conclusions can be drawn from the working principle:

(1) The ZVS of both the leading arm switch tube and the lagging arm switch tube is achieved by utilizing the energy of the secondary output filter inductor. Therefore, the inductance value in series with the primary side can be greatly reduced (even the series inductor is not required, and only the primary leakage inductance of the transformer is used).

(2) When soft switching is implemented, energy is provided by both the secondary inductance and the primary inductance, so ZVS can be achieved over a wider load range.

(3) The conditions for achieving soft switching ZVS by the leading arm switch tube and the lagging arm switch tube are not as stringent as those of the basic circuit, and due to the influence of the secondary side inductance, the difference in soft switching conditions between them is greatly reduced compared with the basic circuit.

3. Converter control circuit design

The control system forms two control loops by collecting the primary bus current and the secondary output voltage: the current inner loop and the voltage outer loop. The principle block diagram is shown in Figure 3-1. UCC3895 is a high-performance current/voltage phase-shift PWM controller produced by TI in the United States. It is an improved version of UC3875 (79); it is most suitable for phase-shift full-bridge circuits, and works with zero voltage switching to achieve local soft switching performance at high frequencies. In addition to the functions of UC3875 (79), its biggest improvement is the addition of an adaptive dead zone setting to adapt to different quasi-resonant soft switching requirements when the load changes. At the same time, because it uses the BCDMOS process, it has lower power consumption and higher operating frequency.

It can be clearly seen from the principle block diagram that the primary bus current is collected through current transformer isolation, and the signal is filtered and slope compensation circuit to obtain the current control signal; and the output voltage signal is adjusted by TL431 and isolated by optocoupler, and then compared with the set voltage reference value to obtain the voltage control signal. After the current and voltage control signals are input into the phase-shifted PWM controller UCC3895, four PWM control signals are obtained through the internal comparator and pulse generation circuit of the chip. However, one thing must be noted, that is, the driving ability of UCC3895 is very weak, so these control signals must be power amplified and isolated before they can drive the switch tubes in the two bridge arms of the main circuit. Among them, the advantage of using the bus current is that it can reflect the penetration of the upper and lower switch tubes in the same bridge arm, thereby providing a certain basis for the protection circuit of the switch tube. In addition, the key to the success of this solution is the slope compensation circuit and the isolation drive circuit.

4. Practical circuit analysis

Figure 4-1 shows the main circuit diagram actually used, in which the filtering and EMI parts mainly consider the processing of series mode and common mode interference. The maximum current flowing through the rectifier bridge is 10A, and a fuse is added to prevent major accidents. _R1 and R2 _form a DC bus voltage detection divider. After the voltage signal is passed through the control and logic circuit, one path is directly given to the solid relay SSR of the bus soft start circuit, and the other path is given to the soft start control circuit SS (Soft Start) part of the control chip to control the soft start of UCC3895, and the delay time between these two soft starts can be adjusted through circuit parameters. _C5 and C6 _are both electrolytic capacitors with a value of 2200uF. CS is a bus current transformer. By detecting the bus current signal and then superimposing it with the Ct terminal voltage signal output by the internal oscillator of the chip through a certain proportion, a slope compensated current signal can be obtained; at the same time, the current detection circuit can also play a pulse by pulse overcurrent protection function, and can prevent the upper and lower tubes of the same bridge arm from being turned on at the same time. Ch is a high-frequency non-inductive _capacitor with a size of 0.033uF. Due to the high operating frequency of the circuit, it is connected as close to the current transformer and the ground as possible in the circuit design. _Q1 - _Q4 are the main switch tubes. The parallel diodes in the figure are their internal equivalent representations, and the capacitors can be external capacitors. Ls is the resonant inductor with a value of 10uH. Tr is the main transformer with a transformation ratio of 1:1. DS1, _DS2_, Lf1 , Lf2 _form the secondary side of the current doubler rectifier. _C7 and C8 _{are
electrolytic capacitors, both of}_which are 2200uF. C9 _is a high-frequency non-inductive capacitor with a size of 1uF.

When the DC voltage is 250V (the load resistance is 10.7Ω and the circuit operating frequency is 100KHz): The voltage across the switch tube G and E (waveform 1) and the voltage across C and E (waveform 2) during soft opening

From the above two figures (a) and (b), it can be seen that: after the voltage drop across the switch tube C and E reaches zero (the anti-parallel diode is turned on before this), the gate drive voltage rises to the gate platform value (about 6V) 100-200ns, and the switch tube begins to conduct at this time, so they are turned on at zero voltage. At the same time, please note that there is a certain difference between the soft opening of the leading bridge arm and the lagging bridge arm. Specifically, the leading arm is easier to achieve soft opening, so its soft opening effect is more obvious under the same conditions.

5. Conclusion

The circuit design is feasible, it combines the advantages of current mode control, phase-shifted PWM control, current doubler rectifier circuit, the latest driver chip and specially designed switching devices:

(1). From the experimental waveform, the leading and lagging bridge arm switching devices of the converter can both achieve zero voltage soft switching very well, and the conditions for achieving zero voltage soft switching and the difference between the two bridge arm soft switching are also smaller than those of the basic circuit. In addition, after adopting the current doubler rectifier circuit, the design of the converter is also simpler: for example, the secondary side of the main transformer only needs a single winding, instead of introducing a center tap like full-wave rectification; and the significant reduction in the secondary side inductance also makes the design of the inductance more convenient.

(2) The use of current mode control can bring a series of benefits. For example, it has advantages that voltage mode control cannot match in preventing transformer core saturation, providing pulse-by-pulse current limiting control, and ensuring the balance of the secondary inductor current of the current doubler rectifier.

(3). High-speed and high-current driver chips make the design of drive circuits simpler and more reliable.

Reference address：Research on Current Mode Controlled Current Doubler Rectifier ZVS PWM Full-Bridge DC-DC Converter

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