A new solution to improve cross regulation of multi-output power supplies

　　Multi-channel switching power supplies are widely used in various complex electronic systems with low power due to their small size and high cost performance. However, with the development of modern electronic systems, the requirements for multi-channel output power supplies are getting higher and higher, such as volume, efficiency, output voltage accuracy, load capacity (output current), cross-regulation rate, ripple and noise. Among them, the cross-regulation rate refers to the rate of change of the output voltage of each channel of the entire power supply when the load current of one channel of the multi-channel output power supply changes, which is an important performance indicator for evaluating multi-channel output power supplies. Affected by the leakage inductance between the windings of the transformer, the resistance of the windings, the parasitic parameters of the current loop, etc., the cross-regulation rate of the multi-channel output power supply has always been the design focus of the multi-channel output switching power supply.

　　At present, the methods for improving the cross-regulation rate can be divided into two categories: passive and active. The active method requires the addition of additional linear or switching voltage regulator circuits. Although a higher cross-regulation rate can be obtained, it is at the expense of the efficiency and cost of the power supply, and is not as good as the passive method in terms of reliability and complexity. When it comes to the passive cross-regulation rate optimization method, experienced engineers will first think of the output voltage weighted feedback control. Secondly, if the flyback circuit is selected, the cross-regulation rate will be further optimized by optimizing the coupling of the transformer windings and optimizing the clamping circuit. If the forward circuit is selected, the output filter inductors of each channel will be coupled together to further optimize the cross-regulation rate. However, when all the above optimization measures have been adopted and still cannot meet the design requirements, usually have no choice but to add a dummy load to exchange efficiency for the cross-regulation rate, or change to a more expensive active optimization design solution.

　　The following is a new TDK-Lambda multi-output solution for improving cross-modulation rate. This solution can further improve the cross-regulation rate using a passive method.

　　As shown in Figure 1, for two output windings with equal number of turns (Ns1=Ns2), we connect a capacitor C1 across the same-name ends of the two jumps, which can greatly improve the cross-regulation rate.

　　Figure 1

　　For the flyback converter shown in FIG1 , considering the leakage inductance of each winding, it can be equivalent to the circuit shown in FIG2 , where Lleak1 , Lleak2 and Lleak3 are the leakage inductances of windings Ns1 , Ns2 and Np respectively.

　　Figure 2

　　Since Ns1=Ns2, Vs1=Vs2 is always maintained during the entire operation of the power supply, so the circuit can be equivalent to that shown in FIG3 , where Is1 and Is2 are the currents flowing through the windings Ns1 and Ns2 respectively.

　　Figure 3

　　When the power supply is working stably, the average voltage across the inductors Lleak1 and Lleak2 is 0V, so the average DC voltage across the capacitor C1 is also 0V. As the capacitance of capacitor C1 increases, the ripple voltage on the capacitor will become smaller and smaller, so Vo1 will become closer and closer to Vo2, that is, the cross-regulation rate of the power supply will become better and better as the capacitance of C1 increases.

　　For the convenience of analysis, we make the following assumptions:

　　1. Ignore the voltage drop of the diode in the circuit and assume that the voltage drop is 0V.

　　2. The capacitance of capacitor C1 is very large, so the resonant period of C1 and leakage inductances Lleak1 and Lleak2 is greater than the switching period of SW1.

　　3. The output voltage of Vo2 is the feedback detection voltage and remains unchanged. The load of Vo2 is heavy, and the load of Vo1 is light. Vo1>Vo2.

　　Based on the above assumptions, the currents of the secondary components during operation of the power supply will be as shown in Figure 4, Is1 and Is2 are the currents flowing through windings Ns1 and Ns2 respectively, Ip is the primary current of the transformer, ID1 and ID2 are the currents flowing through D1 and D2 respectively, and Vc1 is the voltage on capacitor C1.

　　Figure 4

　　Note: This figure only shows the direction of change of voltage and current

　　In order to determine the initial state of the circuit, we take t5 as the beginning of the power supply working cycle. At t5, the current of diode D1 becomes 0, and the voltage Vc1 on capacitor C1 is at its highest value, and:

　　After the diode D1 is turned off, the secondary circuit can be further equivalent to the circuit shown in Figure 5.

　　Figure 5

　　After the primary switch SW1 is closed at time t6, the Vs voltage is sensed as a negative value (as shown in Figure 6). During the period when SW1 is closed, the power supply operates in two stages: the transformer current commutates from the secondary winding to the primary winding (t6~t7) and the transformer stores energy (t7~t9).

　　Figure 6 　　During t6-t7, ID2>0, diode D2 continues to conduct.

　　By the relation

　　It can be seen that both currents Is1 and Is2 drop rapidly until the moment t7 when ID1=Is1+Is2=0, the diode is reversely cut off, and the commutation phase from the secondary winding to the primary winding ends.

　　During the period t7 to t9, the diode D2 is reversely cut off, and the currents Is1 and Is2 are equal in magnitude but opposite in direction.

　　Capacitor C1 resonates with leakage inductance Lleak1+Lleak2 to discharge. Since Is2 is still large after commutation from the secondary side to the primary side of the transformer, Vc1 quickly changes from positive voltage to negative voltage at time t8 and charges in reverse. At the same time, current Is2=-Is1 begins to decrease until SW1 is turned off at time t9 (that is, t0).

　　At t0, SW1 is turned off, and the transformer enters the commutation stage from the primary side to the secondary side. Vs>Vo2>Vo2+Vc1 (at this time Vc1<0), and diode D2 begins to conduct. Currents Is1 and Is2 increase rapidly. At t1, Is1 changes from negative to positive and flows to Vo2 through C1 and D2 (as shown in Figure 7). At t2, the commutation ends. At this time, there is

　　When the transformer primary current is commutated to the secondary current, Vs

　　At t3, the capacitor voltage is charged to Vs=Vc1+Vo2, and as Vc1 increases, Vs

　　Figure 7

　　At t4, diode D1 starts to conduct, and the secondary circuit is equivalent to Figure 3. Current Is1 flows to Vo1 through D1, and the voltage of C1 is clamped at Vc1=Vo1-Vo2, while Is1 continues to decrease. Until t5, Is1=0, diode D1 is reversely cut off, and the power supply completes a switching cycle.

　　Figure 8 shows the average current flow of each branch on the secondary side during the SW1 off period. The average currents of windings Ns1 and Ns2 at the output are:

　　It can be seen from the waveform of Vc1 in Figure 4 that when the switch SW1 is turned off, the voltage Vc1 of the capacitor C1 changes from negative to positive, so Ic1>0. Therefore, it can be concluded that after the capacitor C1 is connected across the windings, when the switch SW1 is turned off, the actual load of the winding Ns1 with a light output load is increased, while the actual load of the winding Ns2 with a heavy output load is reduced, so the cross regulation rate will be improved.

　　Figure 8

　　Currently, this solution has been successfully applied to TDK-lambda's CUT75 series products.

　　Taking CUT75-522 as an example, the power supply usage environment is as follows:

　　Input voltage: 85 ~ 265VAC or 120 ~ 370VDC.

　　Load range: 5V: 0 ~ 8A;

　　+12V: 0 ~ 3A;

　　-12V: 0 ~ 1A。

　　Working temperature: -20 ~ 70℃.

　　By adopting a passive method to successfully achieve the cross regulation rate of +12V and -12V within ±5% by connecting capacitors across the windings. Table 1 below shows the highest and lowest values ​​of each output voltage measured under various output load conditions, as well as the cross regulation rate calculated based on the measured values.

　　Table 1

　　At the same time, because the capacitor is connected across the windings, the CUT75 series power supply can reduce the internal dummy load of the power supply to almost zero while meeting the cross-regulation rate, so the efficiency of the power supply is effectively improved, so that the size of the power supply can be made smaller. The actual measured value of the full-load efficiency of the CUT75 series power supply at an input voltage of 200VAC has reached 85%, which is about 5% higher than similar products on the market, and its size is naturally smaller than similar products on the market.

　　Similar products on the market that can meet the ±5% cross regulation rate mostly use active methods to optimize the cross regulation rate, while the CUT75 series power supply uses a passive method. In comparison, the CUT75 series power supply has more advantages in reliability.

　　CUT75 series power supply physical picture