Single-ended flyback switching power supply design reference

In recent years, with the rapid development of power supply technology, switching power supplies are developing in the direction of miniaturization, high frequency and inheritance, and high-efficiency has been more and more widely used. The converter is particularly suitable for designing low-power switching power supplies due to its many advantages such as simple circuit and efficient DC output.

1 Basic Principle of Flyback Switching Power Supply

The single-ended flyback switching power supply adopts a dual-loop feedback (output DC voltage isolation sampling feedback outer loop and primary coil magnetizing peak current sampling feedback inner loop) control system with good stability. The pulse duty cycle can be quickly adjusted through the PWM (pulse width modulator) of the switching power supply, so as to effectively adjust the output voltage and primary coil magnetizing peak current of the previous cycle in each cycle to achieve the purpose of stabilizing the output voltage. The biggest feature of this feedback control circuit is that it has a faster dynamic response speed, automatically limits the load current, and has a simple compensation circuit when the input voltage and load current change greatly. The flyback circuit is suitable for low-power switching power supplies, and its schematic diagram is shown in Figure 1.

The following is an analysis of the working conditions of current-mode PWM under ideal no-load conditions. Compared with voltage-mode PWM, current-mode PWM adds an inductor current feedback link.

In the figure: A1 is the error amplifier; A2 is the current detection comparator; U2 is the RS trigger; Uf is the feedback sampling of the output voltage Uo, and the feedback sampling and the reference voltage Uref generate the error signal Ue through the error amplifier A1 (this signal is also the comparison clamping voltage of A2).

Assume that the field effect transistor Q1 is turned on, the inductor current iL increases linearly with a slope Ui/L, where L is the primary inductance of T1. The inductor current is sampled on the non-inductive resistor R1 as u1=R1iL. The sampled voltage is sent to the current detection comparator A2 for comparison with Ue from the error amplifier. When u1>Ue, A2 outputs a high level and sends it to the reset terminal of the RS trigger U2. Then the two-input NOR gate U1 outputs a low level and turns off Q1. When the clock outputs a high level, the NOR gate U1 always outputs a low level to block PWM. When the oscillator output clock drops, the two inputs of the NOR gate U1 are both low levels, and Q1 is turned on.

Therefore, from the above analysis, it can be seen that the rising edge of the current-type PWM signal is determined by the falling edge of the oscillator clock signal, while the falling edge of the PWM is determined by the inductor current sink signal and the error signal from the error amplifier. Its working timing is shown in Figure 2.

The main feature of the single-ended flyback switching power supply is the periodic on and off of the main switch tube. When the switch tube is on, energy is continuously stored in the primary coil of the transformer; when the switch tube is off, the transformer uses the inductive energy stored in the primary coil to supply power to the load through the rectifier diode until the next pulse arrives and a new cycle begins.

The pulse transformer in the switching power supply plays a very important role: first, it realizes the conversion of electric field-magnetic field-electric field energy and provides a stable DC voltage for the load; second, it can realize the transformer function. Through the primary winding and multiple secondary windings of the pulse transformer, it can output multiple different DC voltage values ​​to provide DC power for different circuit units; third, it can realize the electrical isolation function of the traditional power transformer, isolate the hot ground from the cold ground, avoid electric shock accidents, and ensure the safety of the user end.

2 Flyback switching power supply design

The most important part of the switching power supply design is the design of the feedback loop. The quality of the feedback loop design directly determines the accuracy and stability of the switching power supply. As mentioned above, the single-ended flyback switching power supply adopts dual-loop feedback. The following will introduce some issues that need to be paid attention to when designing two feedback loops of the switching power supply using the current-mode PWM chip UC3842.

2.1 Output DC voltage isolation sampling feedback external loop

UC3842 is a high-performance fixed-frequency current-mode pulse width integrated control chip designed for offline DC conversion circuits. Its main advantages are that the voltage regulation rate can reach 0.01%, the operating frequency is as high as 500 kHz, the starting current is less than 1 mA, and there are few peripheral components. It is suitable for small switching power supplies of 20 W to 80 W. Its operating temperature is 0 ℃ to 70 ℃, the maximum input voltage is 30 V, the maximum output current is 1 A, and it can drive bipolar power tubes and MOSFETs. UC3842 is packaged in DIP-8 form. Its internal structure block diagram and the functions of each pin can be found in the relevant manual.

The typical application circuit of UC3842 is shown in Figure 3.

DataSheet related to this article:

The working principle of this circuit is: the DC voltage is added to Rin, and then added to pin 7 after voltage reduction, providing the chip with a startup voltage greater than 16 V. When the chip is started, the feedback winding provides the voltage required to maintain the normal operation of the chip. When the output voltage increases, the feedback voltage generated on the feedback winding of the transformer Tl also increases. This voltage is divided into a large voltage-dividing network by R1 and R3, and is sent to pin 2 of UC3842 after voltage division. After being compared with the reference voltage, it is amplified by the error amplifier, so that the duty cycle of the driving pulse of pin 6 of UC3842 is reduced, thereby reducing the output voltage and achieving the purpose of stabilizing the output voltage.

This circuit has a simple structure, is easy to wire, and has low cost. However, the sampling voltage of UC3842 is not obtained from the output terminal, and the output voltage regulation accuracy is not high, so it is only suitable for use in occasions with small loads.

To overcome the above problems, the feedback circuit can be improved by using optocouplers and voltage references for feedback control, which can greatly improve the stability and accuracy. When using this method for feedback control, sampling is required from the output end of the secondary winding. The circuit is shown in Figure 4.

The voltage sampling and feedback circuit is composed of optocoupler PS2701, TL431 and resistor-capacitor network. R5 and C5 in the figure are used for frequency compensation of TL431 and are indispensable. The sampling voltage is obtained by adjusting the voltage divider network composed of R6 and R7. The sampling voltage is compared with the 2.5 V reference voltage provided by the three-terminal adjustable voltage regulator TL431. When the output voltage is normal, the sampling voltage is equal to the 2.5 V voltage reference provided by TT431, then the K-pole potential of TL431 remains unchanged, so the current flowing through the optocoupler U3 diode remains unchanged, and then the current flowing through the optocoupler CE remains unchanged. The feedback potential Uf of pin 2 of UC3842 remains unchanged, then the duty cycle of the output drive of pin 6 remains unchanged, and the output voltage is stable at the set value. When the output 5 V voltage increases for some reason, the output voltage sampling value obtained in the voltage divider network will increase accordingly, so that the K-pole potential of TL431 decreases, the current flowing through the optocoupler diode increases, and then the current flowing through CE increases, so that the potential of pin 2 of UC3842 increases. From the internal schematic diagram of UC3842, we can see that the output voltage Ue of the error amplifier A1 decreases, that is, the clamping voltage of the current detection comparator decreases. Therefore, from Figure 2, we can see that the duty cycle of the output drive of pin 6 of UC3842 decreases, thereby reducing the output voltage, thus completing the feedback voltage regulation process.

2.2 Primary coil magnetization peak current sampling feedback inner loop

The inner loop feedback of the primary coil magnetizing peak current sampling is also a decisive link in the design of the switching power supply. If the inner loop feedback design does not meet the circuit requirements, the switching power supply will not work properly.

When designing the inner loop feedback, a sampling resistor Rs with ground as the reference needs to be connected in series with the switch tube (see R1 in Figure 1 and Figure 4 and R8 in Figure 3) to convert the current of the primary coil into a voltage signal. This voltage is monitored by the current detection comparator A2 and compared with the output level from the error amplifier A1.

Under normal operating conditions, the peak inductor current is controlled by the voltage on Pin 1, where:

Abnormal operating conditions occur when the power supply output is overloaded or output sampling is lost. Under these conditions, the current comparator threshold is internally clamped to 1.0 V.

The maximum peak current of the primary coil of the switching power supply is the maximum current flowing through the primary coil of the transformer during short-circuit protection:

Where: IP is the primary coil inductor current; Pout is the designed output power of the switching power supply; Vin is the input voltage of the switching power supply; D is the duty cycle of the PWM output signal; N is the power efficiency.

According to formula (2) and formula (3), it can be deduced that:

According to the calculated Rs resistance value, the power of the current sampling resistor can be further calculated:

After selecting the current sampling resistor, the sampling signal needs to be sent to the current comparator of UC3842 through an L-type RC low-pass filter network.

From the logarithmic amplitude-frequency characteristics of the low-pass filter, we can see that when the input signal frequency is lower than fh, the output signal is almost exactly the same as the input signal; when the input signal frequency is higher than fh, the output signal will be greatly attenuated.

The signal frequency on the Rs sampling resistor can be measured by an oscilloscope. Therefore, when selecting the RC parameters of the low-pass filter, it is necessary to ensure that the normal sampling voltage on the Rs resistor cannot be attenuated by the filter.

During the design, if the RC parameters are improperly selected, the upper cutoff frequency fh of the filter will be too small, resulting in the attenuation of the normal Rs sampling signal. In this way, when the load increases, the PWM cannot increase the duty cycle of the control pulse, and the transformer will howl due to the heavy load. To solve this problem, the value of the filter capacitor C is reduced, and then fh is increased, so that the normal Rs sampling signal passes through the filter. When the load increases, the switching power supply can stabilize the voltage well, and the transformer howling phenomenon does not occur.

3 Conclusion

The design of switching power supplies is a very practical subject. The method given in this article is only used as a reference. Many practical problems need to be continuously summarized and improved in practice. Only through practice can the design be continuously improved.