Improving the efficiency of medium voltage boost converters in LED TV backlight systems

Low voltage range boost converters are often used in mobile devices to boost the battery voltage (1.2V to 4.2V) to a higher voltage level (e.g., 1.5 to 20V) to power the application circuits. In this voltage range, conduction losses are the main consideration. There are many devices on the market that are specifically designed for these applications, and continuous conduction mode (CCM) is the main operating mode of these devices.

High voltage range boost converters are usually used as PFC converters with 90V to 270VAC input and about 400VDC output. In these applications, conduction loss is not as important as in low voltage boost converters, and switching loss and noise immunity need to be considered more. Therefore, PFC controllers usually adopt some special design elements such as critical conduction (CRM) working mode and higher current sensing voltage. PFC controllers are widely used due to their huge market.

LEDTV backlight applications require a 24VDC input, 180VDC 0.4A output boost converter. Compared to the low-voltage and high-voltage range boost converters mentioned above, this type of medium-voltage boost converter is rarely used in consumer electronics. In this voltage and power rating range, conduction loss, switching loss, and noise immunity all need to be considered, and it is difficult to find a suitable and cheaper device.

Topology and Component Selection Considerations

When designing consumer product solutions, it is always necessary to avoid expensive topologies and components. Moreover, since both the DC input node and the output node (LED array) are located on the secondary side, no isolation is required for the LED backlighting stage. Even though we have other options such as soft-switching resonant half/full-bridge topologies, the boost topology is the best core topology for LEDTV backlighting power supply applications.

Considering that boost controllers for mobile devices have high PWM frequencies (typically 500KHz to 6MHz) and low noise compatibility (voltage mode or low current sensing voltage). PWM controllers for AC/DC power supplies seem to be more suitable because of their high gate drive voltage (over 10V) and high current sensing voltage (typically 0.5V-1.2V). However, most AC/DC PWM controllers operate at frequencies of 50kHz to 100kHz. This frequency range is suitable for power supplies with 90-270VAC input because it balances switching losses and inductor component size. However, for 24VDC input power supplies, the frequency is a bit low because the low operating frequency requires the use of large inductors.

The CRMPFC controller is the best choice because it not only has the advantages of AC/DC PWM controllers (high gate drive voltage and high current sensing voltage), but also can set the operating frequency to the optimal value (200kHz) by selecting the inductor. Even though the feedback loop of the CRMPFC controller works in voltage mode, its sawtooth wave generator and comparator are built into the chip and have a large enough amplitude. Therefore, there will be no problem in terms of noise compatibility.

Improve efficiency

Using a standard CRMPFC controller to implement a boost converter, switching losses are not an issue due to the relatively low input/output voltage and critical conduction mode operation. The issue is the conduction losses. Figure 1 shows the main sources of conduction losses in a boost converter.

We can see that the conduction loss during the conduction period comes from Rsense, Rdson and Rcoil. This article does not discuss how to reduce Rcoil. The following will discuss how to reduce Rsense and Rdson respectively.

In PFC applications, the Rsense value is determined by the maximum rated power. When an abnormal overcurrent condition occurs, the voltage on Rsense should reach the pulse-by-pulse current limit level (Vcslim), and a 10% margin range needs to be retained. Therefore, Rsense can be calculated by the following formula:

For the application discussed in this article, we should also follow this formula. The power consumption of Rsense is:

,thus

We can see that the power consumption of Rsense is proportional to Vcslim. The Vcslim of a standard PFC controller is about 0.5V to 1.2V to avoid false triggering caused by noise. In the FAN7930CM, Vcslim is 0.8V. This value is suitable for PFC applications because the input voltage is relatively high and IQRMS is relatively low. But for 24V input applications, this voltage is too high, making PRsense too large. For example, we use the design tool provided by Fairchild Semiconductor to calculate the power consumption of Rsense for 72WPFC (90VAC input, 400V/0.18A output). We get the result: Rsense=0.289Ω, and the power consumption of Rsense is 0.22W. The efficiency loss on Rsense is then 0.22/72×100%=0.31%. If we use the same design tool to calculate a 72WPFC controller with 24V input and 180V/0.4A output, the result is: Rsense=0.077Ω. The power dissipation in Rsense is 0.96W, so the efficiency loss is 0.96/72×100%=1.33%, which is three times higher than the 90VAC input condition.

[page]In order to reduce the power consumption of Rsense, we designed the "voltage blockup" circuit as shown in Figure 2, using the voltage divider R1 and R2 to introduce a voltage difference between the Vrs and Vsense pins. Through this voltage difference, Vsense can reach Vcslim with a lower Rsense voltage.

In Figure 3, we can see that by adding R1 and R2, Vsense can reach (Vcslim/1.1) even if the voltage drop on Rsense is much lower than Vcslim. This can reduce the power consumption of Rsense. For example, without using R1 and R2, if Rsense is 0.077Ω, when Ipk=10.39A, Vsense is 0.8V. If Vgate=11V, R1=10KΩ, R2=400Ω, Rsense=0.0375Ω, when Ipk=10.39A, Vsense can also reach 0.8V. However, if Rsense=0.0375Ω, the power consumption of Rsense is 0.47W, and the efficiency loss is 0.47/72×100%=0.65%, which is 0.68% higher than using 0.77Ω Rsense.

If Vdss increases, the Rdson of the MOSFET will increase when the MOSFET die size and package are the same. For example, the Rdson of the Fairchild 100VMOS device FDD86102 is 24mΩ. But for the 250VMOS device FQD16N25C with the same package and price, the Rdson is 270mΩ. The conduction loss of the MOSFET device is very different under the conditions of 24mΩ and 270mΩ. We used the same design tool to calculate the conduction loss of the 24VAC input, 180V/0.4A output PFC converter Rdson. The values ​​are 0.9W and 10.08W respectively. Obviously, 270mΩ Rdson is unacceptable. In the standard boost topology, in order to provide an output voltage of 180V, a 250VMOSFET is required to obtain sufficient Vdss margin. In this case, the standard way to reduce the conduction loss is to select a MOSFET device with a lower Rdson. However, under the same Vdss, the MOSFET device with a lower Rdson is not only expensive, but also has a larger Coss. A larger Coss means a larger turn-off loss. Here, we have found another way to reduce conduction loss. This is to use a 100V MOSFET device such as FDD86102 to increase the 24V voltage to 180V. Of course, special methods must be used to solve the voltage problem, such as autotransformers.

Figure 4 shows a boost converter using an autotransformer instead of an inductor. During the on-time, the current flows through the red path just like a standard boost converter, and during the off-time, the current flows through the green path. The voltage on the MOSFET drain is:

If we input N1=3T, N2=7T, Vdiode=1V, Vout=180V, Vin=24V, then Vd is:

Therefore, 100VMOSFET devices can be used.

Design Examples and Test Results

[page]U4，Q35，T3，D36和外部元件构成了这个升压转换器，绕组6-10用于实现零电流检测（ZCD），D37，C42，R39，R40具有两项功能，一项功能是作为箝位线路，吸收N1和N2之间的泄漏电感引起的电压脉冲，另一项功能是监视Q35的漏极电压，反馈至U4的引脚1，实现过电压保护。

Figure 6 is a photo of the top, bottom, and side of the evaluation board. We can see that the addition of R38 improves the efficiency by 1.09%. Figure 7 shows the waveform difference between using and not using the Vrsense voltage booster circuit (R38). Table 2 compares the results with and without the autotransformer. If the autotransformer is not used, D37 should be removed and the cathode of D37 should be connected to 24VVin. We can see that the use of the autotransformer improves the efficiency by 14.06%, and Figure 8 shows the waveform comparison.

Conclusion

标准CRMPFC控制器就其特性、通用性和低价格而言，适用于中等电压升压转换器。传导损耗是其应用的主要挑战。采用电压垫高电路能够降低Rsense所需的峰值电压以期提升转换器的效率。在升压转换器中采用自耦变压器，允许使用低VdssMOSFET器件以减小Rdson，从而显著提升效率。评估电路板的测试结果证实这一思路是可行的。