With an efficiency of up to 99.9%, this control solution protects automotive electronic systems~

ADI has published several publications detailing the ISO 7367-2 and ISO 16750-2 specifications and how to simulate them using LTspice ^{® .}

In its most recent iteration, the ISO 7367-2 electromagnetic compatibility specification focuses on large amplitude (>100 V), short duration (150 ns to 2 ms) transients from relatively high impedance sources (2 Ω to 50 Ω). These voltage spikes can often be eliminated using passive components. Figure 1 shows the ISO 7367-2 Pulse 1 as defined, with the addition of a 330 μF bypass capacitor. The capacitor reduces the spike amplitude from –150 V to –16 V, well within the range supported by the reverse battery protection circuit. ISO 7367-2 Pulses 2a, 3a, and 3b consume much less energy than Pulse 1 and require less suppression capacitance.

Figure 1. ISO 7367-2: Pulse 1 with and without a 330 μF bypass capacitor.

ISO 16750-2 focuses on long pulses from low impedance sources. These transients cannot be easily filtered and typically require an active solution based on a voltage regulator. Some of the more challenging tests include: load dump (test 4.6.4), reverse battery (test 4.7), superimposed alternating voltage testing (test 4.4), and engine start conditions (test 4.6.3). Figure 2 shows a view of these test pulses. The variability of the conditions shown in ISO 16750-2, combined with the voltage and current requirements of the ECU, often requires a combination of these approaches to meet all requirements.

Figure 2. Overview of some of the more stringent ISO 16750-2 tests.

Load dump (ISO 16750-2: Test 4.6.4) is a severe transient overvoltage that simulates a situation where the battery is disconnected but the alternator is supplying a large amount of current. The peak voltage during a load dump is classified as suppressed or unsuppressed, depending on whether avalanche diodes are used at the output of the 3-phase alternator. Suppressed load dump pulses are limited to 35 V, while unsuppressed pulse peaks range from 79 V to 101 V. In either case, recovery may take 400 ms due to the large amount of electromagnetic energy stored in the alternator stator windings. Although most car manufacturers use avalanche diodes, increasing reliability requirements have led some manufacturers to require the peak load dump voltage of the ECU to be close to the voltage in the unsuppressed case.

One solution to the load dump problem is to add a transient voltage suppressor (TVS) diode to locally clamp the ECU supply. A more compact and tighter tolerance approach is to use an active surge stopper, such as the LTC4364, which linearly controls a series N-channel MOSFET to clamp the maximum output voltage to a user-configured level (e.g., 27 V). The surge stopper can help disconnect the output, supports configurable current limit and undervoltage lockout, and can provide the commonly required reverse battery protection using back-to-back NFETs.

For linear regulated power devices, such as surge stoppers, the concern is that the N-channel MOSFET may dissipate significant power when limiting the output voltage during a load dump, or limiting the current during a shorted output. The safe operating area (SOA) limitations of the power MOSFET ultimately limit the maximum current that the surge stopper can source. It also places a limit on how long regulation must be maintained (usually set using a configurable timer pin) before the N-channel MOSFET must turn off to avoid damage. These SOA-induced limitations become more severe as the operating voltage increases, making surge stoppers more difficult to use in 24 V and 48 V systems.

A more scalable approach uses a buck regulator that can operate from a 42 V input, such as the LT8640S. A switching regulator does not have the MOSFET SOA limitations of a linear regulator, but is obviously more complex. The efficiency of a buck regulator allows for high current operation, and its top switch allows the output to be disconnected and allows for current limiting. The buck regulator quiescent current issue has been addressed with the latest generation of devices that consume only a few microamps and maintain regulation under no-load conditions. Switching noise issues have also been greatly improved by using Silent Switcher® technology and spread spectrum techniques.

In addition, some buck regulators can operate at 100% duty cycle, ensuring that the top switch is continuously on, transferring the input voltage to the output through the inductor. In overvoltage or overcurrent conditions, switching action is triggered to limit the output voltage or current respectively. These buck regulators (such as the LTC7862) act as switching surge stoppers, achieving low noise and low loss operation while maintaining the reliability of the switch mode power supply.

A reverse voltage condition (also called a reverse battery condition) occurs when the battery terminals or jumpers are connected in reverse due to operator error. The relevant ISO 16750-2 pulse (Test 4.7) repeatedly applies –14 V to the DUT for 60 seconds. Some manufacturers have added their own dynamic version of this test, initially powering the device (e.g., V IN ₌ 10.8 V) before suddenly applying reverse bias (–4 V).

A quick study of data sheets reveals that few IC designs can tolerate reverse bias, with the absolute minimum pin voltage for ICs typically limited to –0.3 V. Voltages more than a diode below ground can cause excess current to flow through internal junctions, such as ESD protection devices and the body diode of power MOSFETs. Polarized bypass capacitors, such as aluminum electrolytics, can also be damaged under reverse battery conditions.

Schottky diodes can prevent reverse current, but this approach results in greater power dissipation during normal operation when forward current is higher. Figure 3 shows a simple protection scheme based on a series P-channel MOSFET, which can reduce power losses, but may not work smoothly at low input voltages (for example, engine starting) due to the device threshold voltage. A more effective approach is to use an ideal diode controller (such as the LTC4376) to drive a series N-channel MOSFET, which cuts off the input voltage at negative voltages. During normal operation, the ideal diode controller regulates the source-drain voltage of the N-channel MOSFET to 30 mV or less, reducing the forward voltage drop and power dissipation by more than an order of magnitude (compared to a Schottky diode).

Figure 3. Different approaches to solving difficult ISO 16750-2 tests.

The superimposed alternating voltage test (ISO 16750-2: Test 4.4) simulates the effects of the AC output of a vehicle's alternator. As the name implies, a sinusoidal signal is superimposed on the battery rail with a peak-to-peak amplitude of 1 V, 2 V, or 4 V, depending on the severity classification. For all severity levels, the maximum input voltage is 16 V. The sinusoidal frequency is ranged logarithmically from 50 Hz to 25 kHz, then back to 50 Hz over 120 seconds, repeated a total of 5 times.

This test can cause large current and voltage swings in any interconnected filter network that resonate below 25 kHz. It can also cause problems for switching regulators, whose loop bandwidth limitations make it difficult to regulate with high frequency input signals. The solution would be something like an intermediate rectifying element, such as a power Schottky diode, but for reverse voltage protection this is not a good way to solve the problem.

In this case, an ideal diode controller cannot function as well as in a reverse voltage protection application because it cannot switch the N-channel MOSFET fast enough to keep pace with the input. The gate pull-up strength is a limiting factor, typically limited to about 20 μA by the internal charge pump. While the ideal diode controller can turn off the MOSFET quickly, it turns on very slowly, making it unsuitable for rectification except at very low frequencies.

A more appropriate approach is to use the LT8672 active rectifier controller, which can quickly switch N-channel MOSFETs to rectify the input voltage at frequencies up to 100 kHz. An active rectifier controller is an ideal diode controller with two important additions: a large charge reservoir boosted by the input voltage and a powerful gate driver that quickly switches the N-channel MOSFET. This approach can reduce power losses by more than 90% compared to using Schottky diodes. The LT8672 also protects downstream circuitry from reverse battery effects, just like an ideal diode controller.

The engine starting condition (ISO 16750-2: Test 4.6.3) is an extreme undervoltage transient, sometimes referred to as a cold crank pulse, because the worst case battery voltage drop occurs at lower temperatures. Specifically, when the starter engages, the 12 V battery voltage may drop to 8 V, 6 V, 4.5 V, or 3 V, depending on the severity classification (I, IV, II, and III, respectively).

In some systems, a low dropout (LDO) linear regulator or a switching buck regulator is sufficient to support the power rail through these transients, as long as the ECU voltage is below the lowest input voltage. For example, if the highest ECU output voltage is 5 V and it must meet severity level IV (minimum input voltage 6 V), a regulator with a dropout voltage of less than 1 V will suffice. The lowest voltage section of the engine start condition lasts only 15 ms to 20 ms, so a rectifier device (Schottky diode, ideal diode controller, active rectifier controller) after a large bypass capacitor may be able to survive this portion of the pulse if the voltage headroom briefly drops below the regulator dropout voltage.

However, if the ECU must support voltages above the minimum input voltage, a boost regulator is required. A boost regulator can effectively maintain a 12 V output voltage from an input less than 3 V at high current levels. However, there is one problem with boost regulators: the diode path from input to output cannot be disconnected, so naturally the current is not limited at startup or in the event of a short circuit. To prevent current runaway, dedicated boost regulators (such as the LTC3897 controller) integrate a surge stopper front end to support output disconnection and current limiting, as well as reverse voltage protection when back-to-back N-channel MOSFETs are used. This solution can address load dump, engine starting, and reverse battery with a single integrated circuit, but the available current is limited by the SOA of the surge stopper MOSFET.

The 4-switch buck-boost regulator removes this limitation by combining a synchronous buck regulator and a synchronous boost regulator through a shared inductor. This approach can meet the requirements of load dump and engine start test conditions without the current level or pulse duration being limited by the MOSFET SOA, while maintaining the ability to disconnect the output and limit the current.

The switching operation of a buck-boost regulator is determined by the relationship between the input and output voltages. If the input is much higher than the output, the boost top switch is continuously on and the buck power stage steps down the input. Likewise, if the input is much lower than the output, the buck top switch is continuously on and the boost power stage steps up the output. If the input and output are roughly equal (between 10% and 25%), the buck and boost power stages are turned on simultaneously in a staggered fashion. This maximizes the efficiency of each switching region (buck, buck-boost, boost) by limiting switching to only the MOSFETs required to regulate for input voltages above, approximately equal to, or below the output.

Figure 3 summarizes various solutions for dealing with load dump, reverse input voltage, superimposed alternating voltage and engine starting conditions, as well as the advantages and disadvantages of each solution. Several key conclusions can be drawn:

If boost regulation is required, then a 4-switch buck-boost regulator combines the above desired qualities into a single device. It can effectively regulate severe undervoltage and overvoltage transients for extended durations at high current levels. This makes it the most reliable and simple approach from an application perspective, but it also increases its design complexity. However, the typical 4-switch buck-boost regulator has some disadvantages. For one, reverse battery protection is not naturally provided and additional circuitry must be used to address this issue.

The main problem with a 4-switch buck-boost regulator is that a large portion of its operating life is spent in the less efficient and noisier buck-boost switching region. When the input voltage is very close to the output voltage (V _IN ~ V _OUT ), all four N-channel MOSFETs are actively turned on to maintain regulation. As switching losses increase and maximum gate drive current is used, efficiency decreases. Radiated and conducted EMI performance in this region is affected when both the buck and boost power stage hot loops are enabled and the regulator input and output currents are discontinuous.

A 4-switch buck-boost regulator can regulate the large undervoltage and overvoltage transients that occur occasionally, but does so at the expense of high quiescent current, reduced efficiency, and higher noise in the more common, conventional transition region.

The LT8210 is a 4-switch buck-boost DC/DC controller that can operate with a fixed output voltage as usual and supports a new Pass-Thru™ operating mode (Figure 4) that eliminates switching losses and EMI with a configurable input voltage window. The controller operates from 2.8 V to 100 V, regulating the worst battery voltage drop during engine start-up as well as the peak amplitude of an unchecked load dump. It inherently provides –40 V reverse battery protection, which is achieved by adding a single N-channel MOSFET (DG in Figure 5).

Figure 4. A buck-boost controller with pass-through mode solves many of the problems associated with automotive standards testing.

Figure 5. This 3 V to 100 V input buck-boost controller operates with an 8 V to 17 V passband output.

In pass-through mode, when the input voltage is outside the window, the output voltage is regulated to the edge of the voltage window. The top and bottom of the window are configured by the FB2 and FB1 resistor dividers. When the input voltage is within this window, the top switches (A and D) are continuously turned on, passing the input voltage directly to the output. In the non-switching state, the total quiescent current of the LT8210 is reduced to tens of microamps. No switching means no EMI and switching losses, so the efficiency is as high as more than 99.9%.

For those who want the best of both worlds, the LT8210 can be used, which can switch between different operating modes by toggling the MODE1 and MODE2 pins. In other words, the LT8210 can operate as a traditional buck-boost regulator with a fixed output voltage (CCM, DCM, or Burst Mode™) in some cases, and then switch to pass-through mode when application conditions change. This feature is very useful for always-on systems and start-stop applications.

The pass-through solution shown in Figure 5 passes 8 V and 17 V inputs to the output in the window. When the input voltage is above the pass-through window, the LT8210 steps it down to a regulated 17 V output. If the input drops below 8 V, the LT8210 steps the output voltage up to 8 V. If the current exceeds the inductor current limit or the programmed average current limit (via the IMON pin), switching is triggered in the pass-through window as a protection feature to control the current.

Figures 6, 7, and 8 show the LT8210 circuit’s response to load dump, reverse voltage, and startup test conditions, respectively. Figures 9 and 10 show the efficiency improvement achieved and the low current operation that is possible within the pass-through window (the efficiency at low currents is astonishing). Figure 11 shows the dynamic transition between pass-through mode and CCM operation.

Figure 6. Bandpass response to an unsuppressed load dump.

Figure 7. LT8210 response to reverse battery connection.

Figure 8. Bandpass response to an engine cold start.

Figure 9. Efficiency of CCM and passband operation.

Figure 10. No-load input current in pass-through mode (V _{IN = 12 V).}

Figure 11. Dynamic transition between bandpass and CCM operation.

When designing power supplies for automotive electronic systems, the LT8210 4-switch buck-boost DC/DC controller provides an excellent solution with its 2.8 V to 100 V input operating range, built-in reverse battery protection, and its new pass-through operating mode. The pass-through mode improves buck-boost operation, achieving zero switching noise, zero switching losses, and ultra-low quiescent current while regulating the output to a user-configured window level instead of a fixed voltage. The minimum and maximum values ​​of the output voltage are tied to large transients during load dumps and cold cranks, for example, without MOSFET SOA or current or time limitations caused by linear conditions.

The new LT8210 control scheme enables clean and fast transients between different switching regions (boost, buck-boost, buck, and no switching), making it possible to regulate large signals and high frequency AC voltages at the input. The LT8210 can switch and maintain operation between pass-through operation mode and traditional fixed output voltage, buck-boost operation modes (CCM, DCM, or Burst Mode), and the fixed output can be set to any voltage in the pass-through window (for example, V OUT ₌ 12 V in the 8 V to 16 V window). This flexibility enables users to switch between pass-through and conventional buck-boost operation, taking advantage of the low noise, low I _Q , and high efficiency operation of the pass-through mode, and achieving more precise regulation and better transient response in CCM, DCM, or Burst mode.