High-power Ethernet power supply is no longer difficult

　　High Power Ethernet (HPOE) standards have not yet been finalized. Most people expect the standard to be: the supply voltage is increased to 53VDC, the available current per line reaches 750mA, and the total cable impedance does not exceed 12.5 ohms. If someone expects to design for 46VDC (nominal 48V), 720mA, and the cable impedance is 12.5 ohms, that would be the worst case scenario. Because at 720mA, the cable will produce a voltage drop of 9VDC, leaving 37VDC available for operation. This output power at the end of the cable is about 26.6W, and the typical power circuit that follows can only produce a little more than 20W of power. Unfortunately, for many applications, this is still not enough power. One solution is to use multiple Ethernet lines, but this will cause power distribution problems. The following example provides an answer to this problem. The

　　traditional HPOE interface consists of a polarity protection bridge rectifier and a hot-swappable component with an Ethernet power interface. After the HPOE interface is an isolated converter that can provide a stable output. The best case scenario is that these outputs are not load dependent and that they all have good transient response. The typical designs that seemed to contribute at the time used isolated feedback to generate a voltage (typically 5.0VDC) that was then converted to the desired other voltage values. Often, multiple outputs tried to share a feedback loop, but if this was the case, regulation would be more load dependent. In either case, the losses in the bridge rectifier and converter would result in very poor efficiency. Moreover, the isolated feedback loop would also result in very poor transient response. Unfortunately, in the case of HPOE, the whole point is to get more useful power out of as few Ethernet wires as possible without sacrificing performance.

　　Examples are shown of HPOE interfaces and power converters from which we can get a few percent of additional efficiency and provide excellent transient response. Figure 1 shows one of the two HPOE interfaces in a 47W output dual Ethernet pair design. Two N-channel and two P-channel MOSFETs form each bridge rectifier with the lowest loss. Each MOSFET is biased into the ON state by a 150K resistor from the opposite polarity input line. The gate is protected by a low current Zener diode (test current equal to 50uA). Only two MOSFETs with the correct polarity will turn on. The drain-source diodes of the MOSFETs act as a bridge rectifier until the 150K resistor can charge the gate of the MOSFETs. The integrated HPOE interface helps to simplify the circuit and provides all the necessary interfaces and hot-swap functions.

Figure 1: HPOE interface/hot plug

Figure 2: DC-DC converter

　　Figure 2 shows one of the two DC-DC converters. The active clamp forward converter provides very high efficiency and eliminates the need for isolated feedback. The LM5020 active clamp controller has the ability to control the maximum duty cycle. A ramp is generated across the capacitor (C4) that controls the duty cycle. If C4 is connected to the input voltage through a resistor (R2), the duty cycle is inversely proportional to the input voltage and produces a nearly constant output voltage. Fortunately, 1% precision capacitors that do not require feedback and provide excellent regulation are available for only a few cents today. By eliminating these components, the design no longer has any losses caused by various current sensing or limiting. Current limiting in the hot-swap section before the forward converter and in the post regulator at the output provides adequate protection and simplifies the design. A well-regulated Vcc is provided by emulating the regulated secondary-side rectifier/inductor circuit. The LM5025 controller requires only about 10mA, so a large value inductor is required to prevent peak charging because the rectifiers are not synchronous. However, since the current is very small, an inductor with a DC resistance (DCR) of about 32 ohms and a very small package can be used. Linear regulators powered from high input voltages have this function, but the power loss is quite large and the cost is comparable.

　　Although the transformer is a standard 3.3VDC voltage output unit, the feedforward regulation is set to 3.75VDC. When connected in series, it provides a nominal voltage of 7.5VDC, which provides a good intermediate bus voltage for a buck converter or a boost converter. Since the minimum voltage is about 12% higher than the transformer rating, we can easily set the output higher. For a given power situation, this reduces the operating current and saves about 25% of the copper wire losses in the primary and secondary of the transformer.

　　The synchronous rectifier MOSFET requires the selection of the best RDS(on) vs. gate charge value. The MOSFET turns on slowly through the two resistors R15 and R16, and turns off quickly because of D3 and D4 (signal diodes cannot be used). This helps the synchronous rectifier switch at the optimal time. Only one snubber is needed on the side shown by C18 and C12. Pay close attention to the DCR values ​​of all inductors. The DCR value of inductor L3 used on the output is only 4.2 milliohms, but the power loss in its DCR alone accounts for 0.4% of the total system power loss. There are quite a few inductors that can meet the current requirements, and their DCR values ​​are 12 to 16 milliohms. There are only two types of inductors in the entire design that can carry more than a few milliamps of current. The first is the 3.0uH inductor just discussed; the other is the 4.7uH inductor, which always carries less than 2A of current and has a rated DCR value of 9.5 milliohms. Don't lose all or more efficiency by not checking these parameters. The above parameters are also valid for electrolytic capacitors. Aluminum poly capacitors are recommended because of their extremely small equivalent series resistance (ESR).

　　To provide inherent power distribution, the outputs of the two converters need to be connected in series, and Figure 3 shows this configuration. The design used for the test only provides two output voltage values, 5.0VDC and 12.0VDC, but other output voltages can be easily added. If the efficiency of each post-regulator is equally high, the overall efficiency will remain at the same level regardless of the total number of output voltages.

Figure 3: Series configuration at converter output

　　Figure 4 shows a synchronous buck converter that provides 5.0VDC at 7A. This is a typical buck converter in this output range. The MOSFETs used have similar requirements to the active clamp synchronous rectifier, so the same MOSFETs are used. Current sensing is provided by a DCR sense circuit around the inductor. Current sense resistors only waste energy and are quite expensive.

Figure 4: Synchronous buck converter

　　DCR current sensing is limited by the temperature coefficient of the copper wire in the wirewound inductor. R49 and RT1 provide temperature compensation. RT1 should be placed as close to the output of the inductor as possible, and the PCB layout should be designed to keep the thermistor and the inductor winding at the same temperature. The same inductor is used in the active clamp stage. Since the stable Vcc determines the system operating frequency through R41, a bias voltage from the 12V regulator is required to keep Vcc stable. The internal regulator can easily operate at 7.5V or less, and the diode is then connected in parallel with a small voltage multiplier from the 5V output. The bias voltage Vbias should be 8.0V to 15.0VDC.

　　Figure 5 shows a nonsynchronous boost converter that can provide 12.0VDC and 1A. It is a very common design, but there is one point worth mentioning. If the output of the boost converter is shorted, it cannot prevent the short from being applied to the input voltage because there is no in-line switch to prevent it. Sometimes it's a good idea to have a quick-acting fuse to prevent any accidents from happening, as shown in the picture.

Figure 5: Nonsynchronous boost converter

　　The end result is not only excellent performance and price/performance, but also the ability to provide any number of different output voltages. The design shown has been tested to be 87.6% efficient at 37VDC input voltage; under low line input conditions, the design can provide a regulated output of about 47W. This 87.6% is from the Ethernet connector to the regulated output. The actual power stage provides efficiencies well under 90%, whether or not two series conversion stages are used. The lack of isolated feedback also allows the design to be easily customized without having to worry about compensating the stability of the isolated feedback loop. Compensation of the buck and boost back-end regulators is generally easy to implement.

　　This particular example uses a 7.5V intermediate bus. In some cases it may be more appropriate to use a lower bus voltage; in this case, the series connection of the outputs will force the single supply voltage to be very low for best efficiency. In that case it is better to connect the two power stages in parallel, an approach that requires some form of active power sharing between the rails. Of course, it makes no sense to restrict the design to using two supply units in parallel. In fact, there is no theoretical limit to the number of Ethernet cables that can be used for power delivery. An advantage of a parallel configuration is that it provides built-in redundancy, with lower power capabilities if one cable is disconnected. Such circuits have been developed but will not be described here.