Circuit breakers provide overcurrent and precision overvoltage protection

A simple circuit breaker provides sophisticated overvoltage and overcurrent protection.

Requiring only a few inexpensive components, the circuit breaker in Figure 1 responds to overcurrent and overvoltage faults. At the heart of the circuit, an adjustable, high-precision shunt regulator, D2, provides the reference voltage, comparator, and open-collector output, all in a three-pin package.

Figure 2 shows a simplified circuit diagram of the ZR431, D1. The voltage at the reference input is compared to the internal voltage reference VREF, nominally 2.5V. In the power-off state, the reference voltage is 0V, the output transistor is in the off state, and the cathode current is less than 0.1µA. As the reference voltage approaches VREF, the cathode current slowly increases; when the reference voltage exceeds the 2.5V threshold, the device is fully turned on and the cathode voltage drops to approximately 2V. In this case, the impedance between the cathode and the power supply determines the cathode current; the cathode current is in the range of 50µA to 100mA.

Under normal operating conditions, the output transistor of D2 is turned off, and the gate of the P-channel MOSFET (Q4) is turned off through R9 so that the MOSFET is fully enhanced, allowing the load current ILOAD to flow from the power supply –VS through R6 to the load. Q2 and current sensing resistor R6 monitor the magnitude of ILOAD, where the base-emitter voltage VBE of Q2 is ILOAD × R6. For normal values ​​of ILOAD, VBE is lower than the 0.6V voltage required to bias Q2,

In this case the transistor has no effect on the voltage at the junction of R3 and R4. Because the input current to the reference input of D2 is less than 1µA, the voltage drop across R5 is negligible and the reference voltage is actually the voltage across R4.

When ILOAD exceeds the maximum allowed value, an overload condition occurs, and the voltage across R6 increases, causing the base-emitter voltage to be large enough to turn on Q2. As a result, the voltage across R4 and the reference voltage are pulled up to VS, causing the cathode voltage of D2 to drop to approximately 2V. The output transistor of D2 discharges current through R7 and R8, thereby biasing Q3 on. The gate voltage of Q4 effectively controls the power supply through Q3, turning the MOSFET off. At the same time, the source current of Q3 flows through D1 to R4, pulling the voltage across R4, causing the diode voltage to drop below the power supply. Since the base-emitter voltage of Q2 is 0V and it is turned off, no load current flows through R6. The output transistor of D2 latches, and the circuit remains in the fault state, with a load current of 0A. The value of R6 is selected to ensure that the base-emitter voltage of Q2 is less than approximately 0.5V under the condition of the maximum allowed load current.

For overcurrent conditions, the circuit breaker also responds to abnormally large voltages on the power supply. When the load current is within the normal range and Q2 is in the off state, the power supply amplitude and the values ​​of R3 and R4 form a potential divider across the power rails, determining the voltage at the reference input. When the power supply voltage overvoltage condition occurs, the voltage of R4 exceeds the 2.5V reference voltage, and the output transistor of D2 turns on. Once it occurs again, Q3 turns on and the MOSFET (Q4) turns off, effectively isolating the load from the dangerous transient condition.

Now the circuit remains indeterminate until reset. Under these conditions, Q3 controls the gate supply voltage of Q4 to approximately 0V, thereby protecting the MOSFET itself from excessive gate-to-source voltage. Ignoring the negligible voltage value of R5, you can see that the reference voltage is VS × R4 / (R3 + R4). Because the output of D2 goes high when the reference voltage exceeds 2.5V, you can transform the equation to R3 = [(VST / 2.5) – 1] × R4, where VST is the desired supply trip value. For example, if R4 is 10kΩ, a trip voltage of 18V requires R3 to be 62kΩ. R3 and R4 are chosen to set the desired trip voltage value, ensuring that they are large enough so that the potential divider does not overload the supply. Similarly, avoid values ​​that cause errors due to the input reference current.

When you first power up the circuit, you'll find that loads with large inrush currents, such as capacitors, light bulb filaments, and cars, can trip the circuit breaker even though the normal, steady-state operating current is below the level set by R6. One way to solve this problem is to add capacitor C2 to slow down the rate of change of the voltage at the reference input. However, while simple, this approach has a serious disadvantage because it slows the circuit's response time to true overcurrent faults.

Components C1, R1, R2, and Q1 provide another solution. As the voltage increases, C1 initially discharges, causing Q1 to turn on, thereby controlling the reference input to 0V and preventing inrush current from the tripped circuit. C1 then charges through R1 and R2 until Q1 eventually turns off, releasing control of the reference input and allowing the circuit to react quickly to overcurrent transients. With the values ​​of C1, R1, and R2, the circuit allows the inrush current to subside in about 400 milliseconds. Choosing other values ​​will allow the circuit to accommodate inrush currents of any duration appropriate to the load. Once your circuit breaker has tripped, it can be reset by reapplying power or pressing reset switch S1. If your application does not require inrush current protection, simply omit C1, R1, and R2 and connect S1 between the reference input and 0V.

When selecting components, make sure that all components are properly rated for the voltage and current levels they will encounter. There are no special requirements for bipolar transistors, although these transistors, especially Q2 and Q3, should have high current gain, Q4 should have a low resistance value, and the maximum drain-source voltage and gate-source voltage of Q4 must be the same as the highest supply voltage. You can use almost any small-signal diode for D1. As a precaution, if there are very large transient voltages, appropriate Zener diodes D3 and D4 may be necessary to protect D2.

Although this circuit utilizes 431 devices, which are widely available from different manufacturers, not all of them behave exactly the same with respect to D2. For example, testing the Texas Instruments TL431CLP and the Zetex ZR431CLP showed that the cathode current for both devices was 0A when the reference voltage was 0V. However, increasing the reference voltage from 2.2V to 2.45V in steps increased the cathode current from 220 to 380µA for the TL431CLP and from 23 to 28µA for the ZR431CLP—a difference of about 10µA. You must consider the difference in cathode currents when choosing the values ​​of R7 and R8.

The type of D2 you use and the values ​​you choose for R7 and R8 also have an effect on the response time. A test circuit for the TL431CLP, where R7 is 1kΩ and R8 is 4.7kΩ, has a response time of 550ns to a transient overcurrent. Replacing the TL431CLP with a ZR431CLP has a response time of about 1µs. Increasing the values ​​of R7 and R8 to 10 and 47kΩ, respectively, results in a response time of 2.8µs. Note that the larger cathode current produced by the TL431CLP requires correspondingly smaller values ​​of R7 and R8.

To set the overvoltage trip level at 18V, R3 and R4 must have resistance values ​​of 62 and 10kΩ, respectively. Experimental testing of the circuit yielded the following results: D2 uses a TL431CLP and the circuit trips at 17.94V, while D2 uses a ZR431CLP and the trip voltage is 18.01V. The overcurrent detection mechanism, which relies on the base-emitter voltage of Q2, is less accurate than the overvoltage function. However, replacing R6 and Q2 with a high-side current-sense amplifier that generates a ground current proportional to the load current, greatly improves overcurrent detection accuracy. These devices are available from companies such as Linear Technology, Maxim, Texas Instruments, and Zetex.

Circuit breakers prove useful in applications such as automotive systems that require overcurrent detection to prevent erroneous loading and overvoltage protection to shield sensitive circuits from high energy load transients. Except for the small current flowing through R3 and R4, and the cathode current of D2, no current flows out of the circuit under normal, non-tripped conditions to the power supply.

Original English:

Circuit breaker provides overcurrent and precise overvoltage protection

A simple circuit breaker delivers precision overvoltage protection and overcurrent protection.

Anthony H Smith, Scitech, Bedfordshire, England; Edited by Brad Thompson and Fran Granville -- EDN, 6/7/2007

Requiring only a handful of inexpensive components, the circuit breaker in Figure 1 responds to both overcurrent- and overvoltage-fault conditions. At the heart of the circuit, D2, an adjustable, precision, shunt-voltage regulator, provides a voltage reference, comparator, and open-collector output, all integrated into a three-pin package.

Figure 2 shows a simplified view of the ZR431, D1. The voltage appearing at the reference input is compared with the internal voltage reference, VREF, nominally 2.5V. In the off state, when the reference voltage is 0V, the output transistor is off, and the cathode current is less than 0.1 µA. As the reference voltage approaches VREF, the cathode current increases slightly; when the reference voltage exceeds the 2.5V threshold, the device fully switches on, and the cathode voltage falls to approximately 2V. In this condition, the impedance between the cathode and the supply voltage determines the cathode current; the cathode current can range from 50 µA to 100 mA.

Under normal operating conditions, D2’s output transistor is off, and the gate of P-channel MOSFET Q4 goes through R9, such that the MOSFET is fully enhanced, allowing the load current, ILOAD, to flow from the supply voltage, –VS, through R6 into the load. Q2 and current-sense resistor R6 monitor the magnitude of ILOAD, where Q2’s base-emitter voltage, VBE, is ILOAD×R6. For normal values of

ILOAD, VBE is less than the 0.6V necessary to bias Q2 on, such that the transistor has no effect on the voltage at the junction of R3 and R4. Because the input current at D2’s reference input is less than 1 µA, negligible voltage drops across R5, and the reference voltage is effectively equal to the voltage on R4.

In the event of an overload when ILOAD exceeds its maximum permissible value, the increase in voltage across R6 results in sufficient base-emitter voltage to turn on Q2. The voltage on R4 and, hence, the reference voltage now pull up toward VS, causing D2’s cathode voltage to fall to approximately 2V. D2’s output transistor now sinks current through R7 and R8, thus biasing Q3 on. Q4’s gate voltage now effectively clamps to the supply voltage through Q3, and the MOSFET turns off. At the same instant, Q3 sources current into R4 through D1, thereby pulling the voltage on R4 to a diode drop below the supply voltage. Consequently, no load current flows through R6 because Q2, whose base-emitter voltage is now 0V, has turned off. As a result, no load current flows through R6, D2’s output transistor latches on, and the circuit remains in its tripped state in which the load current is 0A. When choosing a value for R6, ensure that Q2’s base-emitter voltage is less than approximately 0.5V at the maximum permissible value of the load current.

As well as responding to overcurrent conditions, the circuit breaker also reacts to an abnormally large value of the supply voltage. When the load current lies within its normal range and Q2 is off, the magnitude of the supply voltage and the values of R3 and R4, which form a potential divider across the supply rails, determine the voltage at the reference input. In the event of an overvoltage at the supply voltage, the voltage on R4 exceeds the 2.5V reference level, and D2’s output transistor turns on. Once again, Q3 turns on, MOSFET Q4 switches off, and the load becomes effectively isolated from the dangerous transient.

The circuit now remains in its tripped state until reset. Under these conditions, Q3 clamps Q4’s gate-source voltage to roughly 0V, thereby protecting the MOSFET itself from excessive gate-source voltages. Ignoring the negligibly small voltage across R5, you can see that the reference voltage is VS×R4/(R3+R4) in volts. Because D2’s output turns on when the reference voltage exceeds 2.5V, you can rearrange the equation as R3=[(VST/2.5)–1]×R4 in ohms, where VST is the required supply-voltage trip level. For example, if R4 has a value of 10 kΩ, a trip voltage of 18V would require R3 to have a value of 62 kΩ. When choosing values for R3 and R4 to set the desired trip voltage, ensure that they are large enough that the potential divider will not excessively load the supply. Similarly, avoid values that could result in errors due to the reference-input current.

When you first apply power to the circuit, you’ll find that capacitive, bulb-filament, motor, and similar loads having large inrush current can trip the circuit breaker, even though their normal, steady-state operating current is below the trip level that R6 sets. One way to eliminate this problem is to add capacitor C2, which slows the rate of change of the voltage at the reference input. However, although simple, this approach has a serious disadvantage in that it slows the circuit’s response time to a genuine overcurrent-fault condition. 　　Components C1, R1, R2, and Q1 provide an alternative solution. On power- up, C1 initially discharges, causing Q1 to turn on, thereby clamping the reference input to 0V and preventing the inrush current from tripping the circuit. C1 then charges through R1 and R2 until Q1 eventually turns off, releasing the clamp at the reference input and allowing the circuit to respond rapidly to overcurrent transients. With the values of C1, R1, and R2, the circuit allows approximately 400 msec for the inrush current to subside. Selecting other values allows the circuit to accommo

date any duration of inrush current you apply to a load. Once you trip the circuit breaker, you can reset it either by cycling the power or by pressing S1, the reset switch, which connects across C1. If your application requires no inrush protection, simply omit C1, R1, R2, and Q1 and connect S1 between the reference input and 0V.

When choosing components, make sure that all parts are properly rated for the voltage and current levels they will encounter. The bipolar transistors have no special requirements, although these transistors, especially Q2 and Q3, should have high current gain, Q4 should have low on-resistance, and Q4’s maximum drain-to-source and gate-to-source voltages must be commensurate with the maximum value of supply voltage. You can use almost any small-signal diode for D1. As a precaution, it may be necessary to fit zener diodes D3 and D4 to protect D2 if extremely large transient voltages are likely.

Although this circuit uses the 431 device, which is widely available from different manufacturers, for D2, not all of these parts behave in exactly the same way. For example, tests on a Texas Instruments TL431CLP and a Zetex ZR431CL reveal that the cathode current is 0A for both devices when the reference voltage is 0V. However, gradually increasing the reference voltage from 2.2 to 2.45V produces a change in cathode current ranging from 220 to 380 µA for the TL431CLP and 23 to 28 µA for the ZR431CL—roughly a factor of 10 difference between the two devices. You must take this difference in the magnitude of the cathode current into account when selecting values for R7 and R8.

The type of device you use for D2 and the values you select for R7 and R8 can also have an effect on response time. A test circuit with a TL431CLP, in which R7 is 1 kΩ and R8 is 4.7 kΩ, responds within 550 nsec to an overcurrent transient. Replacing the TL431CLP with a ZR431CL results in a response time of approximately 1 µsec. Increasing R7 and R8 by an order of magnitude to 10 and 47 kΩ, respectively, produces a response time of 2.8 µsec. Note that the relatively large cathode current of the TL431CLP requires correspondingly small values of R7 and R8.

To set the overvoltage-trip level at 18V, R3 and R4 must have values of 62 and 10 kΩ, respectively. The test circuit then produces the following results: Using a TL431CLP for D2, the circuit trips at 17.94V, and, using a ZR431CL for D2, the trip level is 18.01V. Depending on Q2’s base-emitter voltage, the overcurrent-detection mechanism is less precise than the overvoltage function. However, the overcurrent-detection accuracy greatly improves by replacing R6 and Q2 with a high-side current-sense amplifier that generates a ground-referred current proportional to load current. These devices are available from Linear Technology, Maxim, Texas Instruments, Zetex, and others.

The circuit breaker should prove useful in applications such as automotive systems that require overcurrent detection to protect against faulty loads and that also need overvoltage protection to shield sensitive circuitry from high-energy-load-dump transients. Other than the small current flowing in R3 and R4 and the current in D2’s cathode, the circuit draws no current from the supply in its normal, untripped state.