Study on the resonant pole capacitor snubber circuit of high frequency series inverter-EEWORLD

Collect

Abstract: A lossless resonant pole capacitor buffer suitable for MHz-level high-frequency inverters is discussed. The inverter commutation process is analyzed in detail, the influence of different resonant pole capacitor values on the device turn-off loss and overall loss is studied, and the design method is given. The simulation and experimental waveforms prove the correctness of the theoretical analysis.

Keywords: high-frequency inverter; capacitor snubber circuit; commutation process; lossless

1 Introduction

With the emergence of fast switching devices (such as power MOSFET), the realization of high-frequency induction heating power supply has become possible. The series resonant inverter is the most common topology for realizing high-frequency induction heating power supply. However, if it is operated at a frequency higher than 1MHz, in order to better limit di / dt and du / dt and reduce the switching loss of the device, higher requirements are placed on the inverter's snubber circuit.

Conventional snubbers, such as RCD snubber circuits, use resistors to discharge. As the switching frequency increases, the energy consumed by the snubber also increases, greatly reducing the efficiency of the entire inverter system. However, directly connecting a lossless snubber capacitor in parallel between the drain and source of the MOSFET can effectively reduce the turn-off loss of the switching device and feed back the energy consumed by the resistor in the conventional snubber to the load or power supply, making it more suitable for high-frequency inverter applications. References [1][4] have conducted theoretical analysis and derivation in this regard. On this basis, this paper further explores the characteristics and parameter design of the series resonant inverter containing a resonant pole capacitor snubber at frequencies up to the MHz level, and conducts simulation and experimental verification.

2 Analysis of the commutation process of the inverter with resonant pole capacitor buffer

_{Figure 1 is a simplified main topology circuit of a} series resonant inverter with a resonant pole capacitor buffer. A lossless capacitor is connected in parallel to the drain and source of the switching device MOSFET on the four bridge arms, where _C1 = C2 = C3 ₌ C4 = C. Under inductive load conditions, the switching frequency f should be slightly higher than the resonant frequency f _r , and the phase of the output current i _o lags behind the output voltage U _o_. The specific working process is shown in Figure 2.

Figure 1 Simplified main topology circuit of series inverter with resonant pole capacitor snubber

(a) Before commutation (b) During commutation

Figure 2 Commutation process of a series resonant inverter with resonant pole capacitors

_{State 0} Before commutation, S1 _and S4 _are turned on, and the load current direction is i _{o
>}₀ ; at this time, the voltage on capacitors _C1 and C4 is zero. The voltage on C2 and C3 is U dc _, as _shown in Figure 4( a ).

State 1 S1 _and S4 _are turned off and commutation begins. The load current charges C1 and C4 at i _o / 2 and discharges _through C2 _and C3 , _as shown in Figure 4( _b ).

_{State 2} During the commutation process, when the voltage _on C1 and C4 reaches U _dc , the voltage on C2 and C3 drops to _zero_{,
but the load current has not yet passed zero, it will continue to flow through the internal anti-parallel diodes}_D2 and D3 _, as shown in Figure 4(c).

State 3 After the load current i _{o
passes through zero, S}₂ and S ₃ are turned on, as shown in Figure 4(d).

The above is the working process of the first half cycle. The working process of the second half cycle is similar to that of the first half cycle and is omitted here.

3 Design method of resonant pole capacitor buffer

For a series resonant inverter with a resonant pole capacitor, during operation, if the buffer capacitor has not yet been discharged and the MOSFET device in the same bridge arm is turned on (non-zero voltage turn-on), the capacitor discharge current will directly flow into the switch tube, which will not only cause huge turn-on losses, but also the switch tube is easily damaged by overcurrent. When _fs > 1MHz, the danger of non-zero voltage turn-on is increased.

_In the design, the key is how to determine the value of capacitor C and turn-off angle β0 . A larger C value will reduce the turn-off loss, but at the same time increase the conduction loss; _the smaller β0 , the higher the power factor, but too small β0 will cause the switch tube to turn on at a non-zero voltage. Therefore, when selecting _C and β0 , it is necessary to obtain the smallest possible turn-off loss while ensuring zero voltage turn-on. The following analysis assumes that the quality factor of the load is very _high and the load current is a sine wave.

The waveforms of the output current i _o and the drain-source voltage u _DS of the series resonant inverter are shown in Figure 3. Assuming that i _o changes direction at ω t = 0, and the amplitude of i _o is I _o , then i _o can be expressed as

i _o = I _o sin ωt （1）

Figure 3 Series resonant inverter output current and switch voltage waveforms

At t = - t _{off
, S}₁ and S ₄ are turned off ; at t = - t _on , reverse diodes D ₂ and D ₃ begin to conduct. During the switching period (- t _off < t < - t _on ), C ₁ and C ₄ are charged with 1/2 of the load current i _o , as shown in Figure 2(b). The switch voltage u _{DS
of switch tubes S}₁ and S ₄ can be expressed as

u _DS =(cos ωt -cos β ₀ ) (2)

To ensure zero voltage switching, u _DS must reach U _dc before t = 0. In Figure 3, when ωt = -ξ , u DS _rises to U _dc . Substituting into equation (2) we get

cos β ₀ =cos ξ - (3)

In formula (3) , C , β0 , ω , ξ are all unknown and _it is very difficult to determine their values. Below we first discuss how to choose the value of C.

At the moment when MOSFET is reliably turned off and u _DS rises to U _dc , the load current i _o just drops to zero ( ω t = 0). Assuming C = C _n at this time , we can approximately have

C _n = (4)

Where: ts _is the current fall time.

In the case of zero voltage turn-on, the turn-on loss is close to zero, while the turn-off loss always exists. The resonant pole capacitor connected in parallel at both ends of the switch tube is actually equivalent to a turn-off buffer network. The larger C is, the smaller the turn-off loss is. At the same time, it will also lead to a low power factor and increase reactive power. Usually, the overall loss is minimized near C = 0.45 C _n^[2] . In addition, at high frequencies of MHz, the output capacitance C _oss of the device cannot be ignored. Therefore, the C value can be selected by referring to formula (5).

C = 0.45 C _n － C _oss （5）

Once the C value is determined, β ₀ can be determined by selecting an appropriate ξ according to formula (3) . If the output power is constant, a larger ξ value will result in a larger load current and increase reactive power. Therefore, ξ must be selected as small as possible, assuming ξ = 0. In a series resonant inverter, the switching frequency ω should be slightly greater than the load resonant frequency ω _r so that it works in an inductive state. Considering that physical quantities such as switching frequency and load current will change with the load temperature in actual operation, which may cause the inverter to deviate from the optimal operating point, a certain margin should be left in the selection of β _0. In the design, refer to formula (6) to determine β ₀

β ₀ =cos ^－1 （6）

Where: K should be determined based on the variation range of ω , Io , and _Udc in the actual line , _and is generally taken to be slightly greater than 1.

Based on the above selected values of C and β0 , the design method of other parameters in the series resonant inverter with a series resonant load, such as the inductance angle _φ , the switching frequency ω , and the pulse width tpw of the trigger pulse , is discussed _below .

From Figure 3, it is not difficult to deduce that the expression of _{DC current Id} is ^[1]

I _d = I _o sin ωt d ωt =(cos ξ +cos β ₀ ) (7)

Output Power

P _o = U _dc I _d = U _dc I _o cos ξ - ωCU _dc² (8)

Apparent power S = (9)

Load power factor

PF =≈cos ξ - ωCU _dc (10)

And because the load power factor PF = cos φ (11)

From equations (10) and (11), assuming ξ = 0, we can obtain

cos φ =1－= (12)

The reasonable load inductive angle φ can be determined by formula (12) .

Also because

tan φ = Q (13)

Where: Q is the quality factor.

The switching frequency ω can be determined by equation (13) . It is not difficult to deduce that the optimal pulse width of the trigger pulse at this operating frequency is

t _pw =－〔 t _d(on) ＋ t _r ＋ t _d(off) ＋ t _f 〕(14)

Where: td _(on) , tr , td _(off) , _tf are the internal parameters of _MOSFET .

It can be seen from equation (14) that the selection of pulse width is not only related to β0 and T, but also has a great relationship with the characteristics of the device itself.

4. Impact of resonant pole capacitance on device turn-off loss and overall loss

According to the above analysis, when the inverter works in the best state, its turn-on loss is close to zero, and it is easy to deduce that the energy lost during the turn-off process is

E _off =(15)

The output power factor cos φ is

cos φ =1－(16)

It can be seen from equations (15) and (16) that the larger the C value, the smaller the energy E _off lost during shutdown . However, at the same time, the output power factor cos φ is also reduced. Assuming that the output power remains unchanged, the apparent power will increase, resulting in greater conduction loss.

The following simulation analysis is performed using Pspice software. The switch device is a model established based on the power MOSFET APT10025JVR produced by APT. Its maximum withstand voltage is 1000V, current is 34A, C _oss =1360pF, t _d(on) =22ns, t _r =20ns, t _d(off) =145ns, t _f =16ns. The DC voltage source used is U _dc =100V, the amplitude of the output current is I _o =21A, and the resonant frequency f _r =1MHz. The reference values of the resonant pole buffer capacitance and the turn-off angle are calculated by equations (5) and (6): C =3980pF, β ₀ =34.68°. The corresponding φ =24.33°, f =1.058MHz, t _pw =175ns.

The effect of the resonant pole buffer capacitor on reducing the MOSFET turn-off loss can be seen from the operating waveform, as shown in Figure 4.

(a) C = 0

(b) C =3980pF

Figure 4 Simulation waveform of MOSFET turn-off in series resonant inverter

In the figure: 1—Switch voltage 2—Switch current 3—Turn-off power loss

The following simulation analysis is performed on the impact of different buffer capacitor values on the device shutdown power consumption and average loss. The simulation results are listed in Table 1.

Table 1 Comparison of the impact of different buffer capacitance values on device turn-off loss and average loss

C /pF	Turn-off loss/μJ	Average loss/W
0	31.88	37.7
2000	21.15	31.8
3000	17.92	31.1
3980	15.35	30.5
4500	14.77	32.5
6000	14.58	34.6

It can be seen from Table 1 that when C = 3980pF, β ₀ = 34.68°, the switch device works in the zero voltage turn-on state, and the overall loss is acceptable. If the capacitance value is too small, the turn-off loss is particularly large; if the capacitance value is too large, on the one hand, its effect of reducing the turn-off loss is significantly reduced, and on the other hand, it will also cause huge conduction loss.

5 Experimental Results

In actual high-frequency and high-power series resonant circuits, it is quite difficult to measure the turn-on and turn-off losses of the power device MOSFET. Due to practical conditions, the single-tube test circuit with inductive load shown in Figure 5 is used in the experiment. The power MOSFET device selected is IXFX24N100 produced by IXYS. C _oss =750pF, t _d(on) =35ns, t _r =35ns, t _d(off) =75ns, t _f =21ns. The DC voltage is U _dc =100V output by three-phase rectification . The switching frequency f =1.005MHz. Because the test circuit does not constitute a series resonant inverter, there is no need to consider the influence of the turn-off angle β _0. The experimental waveform is shown in Figure 6.

Figure 5 Simplified single-transistor test circuit with inductive load

(a) MOSFET operating waveform when C = 0

(b) When C = 0, the operating waveform of the MOSFET at the time of shutdown

(d) When C = 500PF, the working waveform of MOSFET at the time of shutdown

Figure 6 Operating waveform of MOSFET in the test circuit

(CH1 is the switching current waveform, CH2 is the switching voltage waveform after the oscilloscope probe is attenuated by 10)

6 Conclusion

1) In a series resonant inverter with a frequency up to MHz, a lossless capacitor of appropriate size is connected in parallel across the drain and source of the switching device to reduce the turn-off loss;

2) The larger the resonant capacitor value, the smaller the turn-off loss, but the overall loss increases. A compromise should be considered when selecting the C value;

3) In actual operation, as the load temperature increases, the inverter deviates from the optimal operating point. The selection of parameters should leave a certain margin to ensure that the MOSFET device in the same bridge arm is turned on only after the buffer capacitor is fully discharged to achieve zero voltage turn-on.

Reference address：Study on the resonant pole capacitor snubber circuit of high frequency series inverter

Previous article：A brief discussion on the technical core of electrodeless lamp ballast (inverter)
Next article：Design of a Unipolar Frequency-doubling Voltage-type SPWM Soft-Switching DC/AC Inverter

Popular Resources
Popular amplifiers