Development process of universal IGBT variable frequency power supply

1 Introduction

At present, some equipment still uses the traditional 400Hz variable frequency unit for power supply, which has the disadvantages of being bulky, inefficient, noisy, poor dynamic quality, and poor output waveform. It is an inevitable trend to replace it with a static variable frequency power supply. Although the early thyristor static variable frequency power supply has overcome many shortcomings of the variable frequency unit, the shutdown of the thyristor depends on the load or the additional shutdown circuit, the control is complex, the dynamic performance is not ideal, and it is difficult to make new breakthroughs in technical performance. The variable frequency power supply proposed in this paper fundamentally overcomes the above disadvantages and is a static variable frequency power supply with excellent performance.

2 Main circuit and system control structure

2.1 Main circuit structure of variable frequency power supply

The main circuit structure is shown in Figure 1. JS is a soft start control to avoid the impact of surge current on the rectifier module when powering on.

The SPWM control strategy, which is popular in the industry, is adopted. Due to the high frequency of the carrier frequency, the center of the first group of harmonics of the SPWM pulse wave migrates to the high frequency end, far away from the fundamental frequency, as
shown in Figure 2. This makes the output filter network small and lightweight, and the dynamic quality is also improved.

The output filter network uses a two-element K-type low-pass filter [5]. The parameters of the filter elements L and C are selected as follows:

L = R/ (πf c) (1)

C = 1/ (πf c R) (2)

Where fc is the high cutoff frequency of the passband and
R is the characteristic impedance of the filter.

2.2 System control structure

The system control structure is shown in Figure 3. As an auxiliary loop of the output voltage control loop, the current loop can successfully limit the output current of the inverter to prevent the inverter from overloading and improve the system stability.

In the figure, Uge is the voltage setting,
Upc is the bias correction,
Igd is the current setting,
and Uxl is the current limiting setting.

3 Holding effect and protection technology[1,2]

3.1 Holding effect

IGBT is composed of four layers of PNPN, and a parasitic thyristor is formed inside, which may be locked due to regeneration. There are two models of IGBT locking effect: static
locking during stable conduction and dynamic locking during shutdown.

3.2 Static holding effect

The equivalent circuit of IGBT is shown in Figure 4a. α1 and α2 are the current amplification factors of VT1 and VT2 respectively and are functions of voltage and current. If α1 increases, the hole current Ih through the P base region also increases. When Up = Ih Rp > 0.7V, the NPN tube is turned on and positive feedback occurs between VT1 and VT2. It is known that when α1 + α2 = 1, the IGBT is locked and the gate loses its control function. The IGBT will be destructively damaged.

3.3 Dynamic holding effect

The equivalent circuit considering the junction capacitance is shown in Figure 4b. When the IGBT is turned off, the J2 junction bears almost all the high voltage due to reverse bias. The junction capacitance Cj2 has the greatest impact, and only the impact of Cj2 is considered. Re-adding
dv/dt causes Cj2 to generate a displacement current iDis:
iDis∝dv/dt (3)

At this time, it should be dynamic α. If the change of α with voltage is not considered and only the influence of current on it is considered, the dynamic αs is defined as:

It can be seen from the above formula that when the lock occurs, αs1 + αs2 = 1. At this time, it has nothing to do with the static α1 and α2. The tube current increases rapidly with the iDis displacement current, and it is most harmful to re-apply dv/dt during shutdown.

3.4 Protection against the holding effect

From the above, we can see that the holding effect of IGBT is determined by the special structure of the device. A good peripheral circuit should be designed for IGBT to suppress the occurrence of holding, mainly from the following aspects.

(1) Prevent IGBT from exceeding thermal limits

The holding current of IGBT is related to the temperature, see Figure 5. The temperature of the heat sink should not exceed 70°C. As the temperature rises, the bias voltage of the NPN tube is no longer 0.7V, but decreases with the increase of temperature; the lateral resistance RP of the p+ region increases with the increase of temperature, and the influence of both causes the holding current to decrease.

(2) Select reasonable driving conditions

The dynamic and static characteristics of IGBT are closely related to the gate drive conditions. The forward and reverse drive voltage ±Uge and the gate resistance Rg have different degrees of influence on the saturation voltage drop, switching loss, short-circuit tolerance, etc. of IGBT. Experience shows that the forward drive is preferably 13V ≤Ug ≤15V, and the reverse drive is preferably -7V ≤-Uge ≤-5V. If the switching loss is allowed, Rge should be appropriately selected.
(3) Use a buffer circuit to limit overvoltage

The surge voltage generated when the IGBT is inductively turned off may cause the IGBT's turn-off trajectory to be outside the safe operating area. On the other hand, it increases the tube power consumption and the temperature rise is not conducive to the suppression of the switch
. A buffer circuit must be used to eliminate this switching surge. The buffer circuit adopts a discharge-blocking structure, as shown in Figure 6. The parameters are selected according to the following relationship:

Where Ls is the lead inductance, measured in 1μH/m

Io ———IGBT maximum pulse current value

K——Rated reduction factor, K=1 for non-repetitive, K=0.8 for repetitive

Ucep——Peak voltage between collector and emitter, Ucep = Ud + UFM + Lsd i/d

Ud ———DC high voltage

UFM - Transient forward voltage drop of diode, 40~60V for 1200V level

The measured voltage peak ΔU = Ucep - Ud <100V, the buffering effect is quite obvious.

(4) In case of overcurrent or short circuit, the IGBT should be turned off slowly.

In the event of a fault, as the MOSFET channel decreases during shutdown, current will flow through Rp, causing Up to increase, and the IGBT may enter a latching state. Simply turning off the IGBT quickly will produce
larger di/dt and dv/dt, which may also cause the IGBT to enter a latching state. It should be tried to mitigate the fault under the premise that the IGBT tolerance allows.

(5) Reasonable selection of device grade and switching frequency

The current level of the IGBT power module is selected according to the following formula:

ic = NPo/ (ηDmaxUdmin) (14)

Where Po is the rated output power,
N is the power margin factor,
η is the efficiency
, Dmax is the maximum duty cycle,
Udmin is the minimum DC high voltage.

The preferred frequency range of high-speed IGBT is 10-15kHz (hard switching). If the switching frequency is too high, the tube power consumption will be large, the temperature will rise, and the reliability will decrease. Taking the single-phase 4kW static variable frequency power supply as an example,
Fuji 2MBI50L-120 power module is selected. The frequency modulation ratio mf = 33, the carrier frequency f = 400 × 33 = 13. 2kHz. After the main circuit multiplication, the frequency of the SPWM pulse wave output by the inverter bridge is 26. 4kHz, and its spectrum is shown in Figure 2.

4 Control, drive and protection circuits

The modulation wave generation circuit composed of EPROM and D/A is currently a better method. The reference sine is calculated offline according to the regular sampling method and stored in the EPROM. If it is a three-phase power supply,
the three phases of the reference sine are 120° apart, and a common EPROM with the minimum capacity can be used. The basic circuit is shown in Figure 7.

Because the hypotenuse of the triangular wave carrier intersects with the reference sine wave at the step, there is no jitter problem in the comparator, and no additional measures are required, which is stable and reliable.

5 Main technical indicators

Single-phase 4kW variable frequency power supply:

Input 50Hz, 380V, ±10%

Output single phase 115V/230V, ±10% adjustable

Voltage regulation < 1 %

Output waveform: sine wave, THD < 3%, single harmonic < 2%

Frequency 400Hz, ±30Hz adjustable

Overload capacity 120%, 10 minutes

Efficiency > 80 %

6 Conclusion

The single-phase 4kW variable frequency power supply has been successfully tested on two radars. Based on this, power supplies of various specifications have been derived. It has achieved practical application and serialization. It has been widely used in the military, colleges, civil aviation, scientific research institutions, etc., and users have given good feedback.