Using Signal Conditioning ICs to Drive Strain Gauge Bridge Sensors

Strain gauges are available with a wide range of zero-strain resistors, and a wide range of sensor materials and related technologies, but a few values ​​(e.g., 120Ω and 350Ω) are used in a wide variety of applications. In the past, standard values ​​were easily interfaced to basic magnetic reflectometers that included matching input impedance networks, simplifying strain measurements.

Types and composition of strain gauges

Metal strain gauges are produced using a number of alloys, selected to minimize the difference in temperature coefficient between the strain gauge and the strain gauge material. Steel, stainless steel and aluminum have become the main sensor materials. Beryllium copper, cast iron and titanium are also available, and "mostly" alloys have driven the high-volume, low-cost production of temperature-compatible strain gauges. 350Ω copper-nickel alloy strain gauges are the most commonly used.

Thick and thin film strain gauges are reliable and easy to produce for the automotive industry. They are usually made of ceramic or metal substrates with insulating materials deposited on the surface. The strain gauge material is deposited on the surface of the insulating layer by vapor deposition. The sensing sheet and connecting wires are engraved on the material using laser vaporization or photomask and chemical etching techniques. Sometimes a protective insulating layer is added to protect the strain gauge and connecting wires.

Strain gauge materials typically include specialized alloys to produce the desired strain gauge impedance, resistance to pressure changes, and (for temperature stability) the best temperature coefficient match between the sensor and the base metal. Strain gauge and bridge resistors with nominal 3kΩ to 30kΩ have been developed for this technology to produce pressure and force sensors.
Bridge Excitation Technology
Strain gauges, thin film and thick film strain gauge sensors typically use a Wheatstone bridge. The Wheatstone bridge converts the resistance generated by the strain of the strain gauge into a differential voltage (Figure 1). When the excitation voltage is applied to the +Exc and -Exc terminals, a differential voltage proportional to the strain appears on the +VOUT and -VOUT terminals.

Figure 1. Strain gauges connected in a Wheatstone bridge configuration.

In a semi-active Wheatstone bridge circuit (Figure 2), only two of the bridge elements are strain gauges, which respond to strain in the material. The output signal of this configuration (typically 1mV/V at full-scale load) is half that of a fully active bridge.

Figure 2. Strain gauges connected in a semi-active Wheatstone bridge configuration.

Another fully active bridge circuit (Figure 3) uses more than four active 350Ω strain gauges. The characteristic bridge resistance is 350Ω, the output sensitivity is 2mV/V, and the strain gauges use strain materials over a large range.

Figure 3. A 16-gauge Wheatstone bridge configuration
The Effect of Temperature on Sensor Performance
Temperature causes a shift in the zero-load output voltage (also called offset) and a change in sensitivity (also defined as full-scale output voltage) under load, which can adversely affect sensor performance. Sensor manufacturers introduce temperature-sensitive resistors into the circuit to compensate for the first-order effects of these changes, as shown in Figures 1 to 3.

As temperature changes, resistors RFSOTC and RFSOTC_SHUNT modulate the bridge excitation voltage. Generally, RFSOTC materials have a positive temperature coefficient and the bridge excitation voltage decreases as temperature increases. As temperature increases, the sensor output becomes increasingly sensitive to load. Reducing the bridge excitation voltage reduces the sensor output, effectively canceling out the inherent temperature effects. Resistor RSHUNT is insensitive to temperature or strain and is used to adjust the amount of TC compensation produced by RFSOTC. An RSHUNT of 0Ω cancels out all effects of RFSOTC, while an infinite value (open circuit) enables all effects of RFSOTC. This method works very well to compensate for first-order temperature sensitivity, but does not address more complex higher-order nonlinear effects.

Temperature compensation of offset variation is accomplished by inserting temperature sensitive resistors in one arm of the bridge. These resistors are ROTC_POS and ROTC_NEG shown in Figures 1 to 3. The shunt resistor (ROTC_SHUNT) adjusts the amount of temperature effect introduced by ROTC_POS or ROTC_NEG. The use of ROTC_POS or ROTC_NEG depends on whether the offset has a positive or negative temperature coefficient.
How to implement current excitation drive
Using current to excite bridge sensors is difficult due to the variation of bridge resistance with load and excessive or reverse current in the built-in sensitivity compensation network (RFSOTC and RFSOTC-SHUNT shown in Figure 2).

Various methods can be used to solve these problems and achieve current excitation drive. A simple method is to use the MAX1452 and configure it to achieve voltage drive. The circuit includes very few external components, which can meet the high current requirements required for voltage excitation. The MAX1452 is a fully integrated signal conditioning IC that completes sensor excitation, signal filtering and amplification, temperature linearization of offset and sensitivity, etc.

The MAX1452 is designed primarily for silicon piezoresistive transducers (PRTs) in pressure sensing. It uses four 16-bit Σ-Δ DACs (DA converters), a temperature sensor, and a temperature coefficient table to complete temperature compensation and linearization of the bridge sensor (Figure 4). Temperature compensation and amplification are accomplished through an analog signal path between the sensing unit and the voltage output. With very little external circuitry, the IC can accommodate metal sheets or thick-film strain gauges, providing voltage excitation and stronger current drive capabilities for the Wheatstone bridge.

Figure 4. The MAX1452 is a fully integrated signal-conditioning IC for bridge sensors.

The MAX1452 includes a PRT current excitation circuit (Figure 5). The circuit includes a current mirror (T1 and T2) that amplifies the small reference current by a factor of 14, enough to drive a 2kΩ to 5kΩ PRT sensor. The reference current is obtained by applying a voltage to RISRC and RSTC. This voltage is set by a 16-bit precision full-scale output D/A converter (FSO DAC) in the feedback loop of op amp U1.

Figure 5. PRT bridge excitation circuit diagram

The FSO DAC uses a Σ-Δ architecture, with digital inputs from a temperature coefficient table in flash memory. A unique 16-bit coefficient is provided to the DAC every 4ms for every 1.5°C temperature increment. The DAC output voltage drives the gate of p-channel MOSFET T1, which in turn drives sufficient current to RISRC and RSTC to produce a voltage equal to the FSO DAC voltage. The current through T1 is mirrored by T2 and amplified 14 times to become the bridge drive current.

Resistor RSTC enables first-order modulation of the sensor excitation current as a function of temperature. For silicon PRT transducers, the temperature is obtained from the resulting sensor bridge voltage when current is passed through the bridge. These sensors have a very good transfer function between the bridge resistance and temperature. By exciting the bridge with current, you can adjust the resulting bridge voltage and use it to perform first-order compensation for offset and sensitivity. This can be done by connecting the bridge voltage (pin BDR) to the reference input of the full-scale output temperature compensation DAC (FSOTC DAC). However, keep in mind that current excitation is generally not suitable when using metal or thick-film strain gauges.
Voltage Drive Circuit
The MAX1452's internal 75kΩ resistor can be used as RISRC and RSTC, or an external resistor can be connected through switches SW1 and SW2, as shown in Figure 5. The ISRC pin accesses the op amp to achieve voltage feedback for the bridge drive. Figures 6, 7, and 8 show three different voltage drive circuits.

Figure 6. High impedance sensor circuit diagram, no external components used.

Figure 7. Low impedance sensor circuit diagram with npn transistor

Figure 8. Circuit using external RSUPP driver

For high impedance sensors above 2kΩ, the simple circuit in Figure 6 provides voltage drive excitation for the bridge. Opening SW1 and SW2 disables the FSOTC DAC modulation circuit. Connecting pins ISRC and BDR forms an op amp feedback loop to obtain bridge excitation voltage feedback. By sourcing current into the bridge, transistors T1 and T2 (in parallel) increase the bridge voltage to equal the FSO DAC voltage.

Low impedance (120Ω to 2kΩ) strain gauges or thick film resistors connected in a Wheatstone bridge circuit cannot be driven directly by T2. An external npn transistor in an emitter-follower configuration solves this problem (Figure 7). The current through the npn transistor comes directly from the collector VDD supply. Driving T1 and T2 enough to conduct turns on the npn transistor, allowing op amp U1 to increase the bridge voltage. To close the loop, the bridge voltage at ISRC is fed back to the op amp. The bridge voltage is regulated to match the FSO DAC output voltage, and a small 0.1µF capacitor is added across the bridge to maintain stability.

The base-emitter voltage (VBE) of the npn transistor has a large temperature coefficient, and the effect of this term in the equation is eliminated through the feedback of U1. At low temperatures, VBE is large, and the maximum bridge voltage is limited to:

VBRIDGEMAX = VDD - VT2SAT - VBE

Similar to VBE temperature compensation, the control feedback loop eliminates the TNPN gain temperature component from the equation.

Another way to provide enough drive current for a low impedance bridge is to add a small external resistor in parallel with T2 (RSUPP in Figure 8). RSUPP ensures that the bridge voltage is slightly less than the required value (3.0V for VDD = 5.0V). T2 provides more current, raising the bridge voltage to the required value. Since T2 provides the minimum current when it is in the OFF state, RSUPP should be adjusted for the worst-case small bridge voltage. Similarly, the maximum current capability of T2 (2mA at VBDR = 4.0V) determines the maximum bridge voltage modulation available. This circuit can be used for bridge sensors with relatively low temperature coefficient of sensitivity (TCS), which do not require large bridge voltage modulation.

U1 feedback eliminates the sensitivity effect caused by the RSUPP temperature coefficient. When designing the circuit, the maximum and minimum RSUPP power derating should be considered to ensure appropriate drive current margin.

Summarize

The MAX1452's flexible bridge excitation method greatly increases the user's design freedom. This article focuses on voltage drive circuits with and without current amplification, and introduces other bridge drive configurations. Other design considerations include using an external temperature sensor in the control loop, feeding the OUT signal into the loop, and implementing sensor linearization (i.e., nonlinearity with respect to the measured parameter).