Single-ended active clamp DC/DC converter

    Abstract: The performance of single-ended forward and flyback converters is greatly enhanced by the use of active clamping/recovery technology. Its advantages are high efficiency, low external interference and device stress. This article introduces two active clamp circuit topologies, analyzes the circuit working status, and derives the relationship between magnetizing current and load current. Finally, the advantages of this active clamped DC/DC converter are summarized.

    Keywords: power conversion switching power supply ZVS

1 Introduction

    In a switching power supply, DC voltage can be converted from one level to another. Such as buck circuit, boost circuit and buck-boost circuit. However, when it is required to convert a relatively high DC voltage to a relatively low DC voltage, the efficiency of conventional conversion technology is low. Especially when the operating frequency of the converter is above 1MHz, the switching loss becomes particularly large.

    The circuit in Figure 1 is a conventional buck (forward) converter. During normal operation of the converter, the switch S1 is turned on, adding the difference between the input voltage and the output voltage to the inductor L1, causing the current in the inductor L1 to increase and charging the input capacitor CS; this current is then sent to the load RL. superior. When the switch S1 is turned off, the polarity of the inductor L1 is reversed, causing the diode D1 to conduct. Then, the current flows through D1 and L1, and its amplitude gradually decreases until S1 is turned on again, and the next working cycle begins.

    Figure 2 shows a conventional buckboost converter, which uses a transformer T to isolate the input and output voltages. The converter can make the amplitude of the output voltage greater or smaller than the amplitude of its input voltage. The disadvantage of this circuit is that the switch current and diode current are both larger than those of a basic buck or boost converter.

    The DC/DC power converter with active clamp introduced in this article can operate with zero-voltage resonant conversion at a switching frequency above 1MHz. In the circuit, only one magnetic core is needed to act as both inductor and transformer. By changing the turns ratio, the required voltage is obtained. The control of its output characteristics is the same as that of ordinary converter topology. Using zero-voltage resonant conversion and transformer isolation technology, there are no special requirements for the magnetic core. The control part of this circuit adopts pulse width modulation technology (PWM), which has high operating frequency, high efficiency, and input and output isolation.

2 Circuit structure description

    Figure 3 shows the DC/DC converter circuit with active clamping that this article focuses on. The circuit uses three switching tubes S1, S2 and S3, a transformer T, and the filter capacitors on the primary side and secondary side of the transformer are Ci and Cs respectively. For the convenience of analysis, it is assumed that the capacitance is large enough and the capacitance voltage is a constant value during the entire switching cycle; the coupling coefficient of the primary and secondary windings of the transformer is 1; the switching tube is ideal, that is, there is no power consumption, and it can pass the switch in either forward or reverse direction. current. In addition, in the analysis, only the single output form is considered. To output several voltages, the secondary winding can be added.

    Usually, an ordinary timing circuit (not shown) is used to control the work of the three switching tubes. The control waveform is shown in Figure 4. During operation, the active clamp switch S1 and the synchronous switch S3 are driven by the same signal ug (on at the same time and off at the same time), as shown in the waveform of Figure 4(a). S2 is driven with the opposite signal. In this way, when S1 and S3 are on, S2 is off, and vice versa. Because it is assumed that S1, S2, and S3 are all ideal switching tubes, that is, the turn-on and turn-off are completed instantaneously. In fact, the switching time is between 30ns and 120ns, and is generally driven by a waveform that turns off first and then turns on.

3 Circuit working status analysis

    Figures 5 and 6 show the two working states of the circuit in Figure 3. Assume that the circuit is already operating in a steady state at the beginning, as shown in Figure 5, S2 is turned on, the current in the primary winding of the transformer increases, charging the capacitor CP, and the output current Io is completely supported by the capacitor CS. In the state shown in Figure 6, S1 and S3 are turned on. This allows the energy stored in the capacitor CP and the inductor LP to be transferred from the primary side of the transformer to the secondary side load.

    The working cycle of S2 is T, the duty cycle is D, and the conduction interval is part of the working cycle, that is, DT. The conduction time interval of S1 and S3 is T-DT=T(1-D). During the period T, the average value of the voltage across the primary winding is zero, that is

    (U i -nU o )DT - nU o (1 - D)T=0 (1)

    U i D=nU o     (2)

    D=nU o /U i    (3)

    In the formula, n is the turns ratio of the transformer. Equation (1) is shown in Figure 4(b). Likewise the average current in Cs is zero. When S2 is turned on, Cs supplies the load current Io. When S1 and S3 are turned on, Cs is charged to compensate for the energy output by Cs when S2 is turned on. Under ideal conditions, the current ICS in Cs can be considered to be basically rectangular, as shown in Figure 4(c). When S2 is turned on, the Cs input current ICS and the output current Io are equal in amplitude and opposite in phase, that is,

    I CS =-I o    (4)

    During the conduction period of S1 and S3, the input current ICS of Cs is equal to the difference between the current Is in the secondary winding and the output current Io, that is

    I CS =I s -I o    (5)

    Because the average current on the capacitor CS is zero, then

    -DIo+(1-D)(IS-Io)=0 (6)

    The current Is in the secondary winding can be expressed as

    Is=Io/（1－D） (7)

    During the conduction period of S1 and S3

    ICS=Io/(1-D)-Io (8)

    =Io·D/(1－D) (9)

    Substitute equation (3) into equation (9) to get

    ICS=Io·nUo/（Ui－nUo） (10)

    The input current ICS in Cs is shown in Figure 4(c), the output current Io is shown in Figure 4(d), and Is is shown in Figure 4(e).

    According to linear superposition, the current in the primary winding of the transformer consists of three parts: the first part is the magnetizing current ILpm, which is caused by the voltage Ui adds to both ends of the primary winding when S2 is turned on. It has nothing to do with the output current; the second part is the current It is during the conduction period of S1 and S3 that the current in the secondary winding induces the current in the primary winding, represented by ILP1-3; the third part of the current is generated by the input current ILP2 during the conduction period of S2.

    The magnetizing current is determined by the voltage applied to the primary winding, the winding inductance, the switching period T and the duty cycle D. When S2 is turned on

    iLPm(t)=ILPm(t0)+(Ui / Lp)t (11)

    During the conduction period of S2, the peak-to-peak magnetizing current is:

    (ILPm)pp=(Ui-nU0 / Lp)DT (12)

The peak-to-peak current during the conduction period of     S 1 and S 3

can be calculated by the same method     (ILPm)pp=(nU0 / Lp)(1-D)T (13)

    Under steady-state conditions, equation (12) is equal to equation (13).

During the conduction period of     S 1 and S 3

, the current ILP1-3 generated by the load current on the primary side can be borrowed from the turns ratio relationship of the transformer to reflect equation (7) to the primary side to obtain     I LP1-3 = (I s / n)=I o / n(1-D) (14)

    During the conduction of S2, the current ILP2 generated by the load current on the primary side can be considered like this: During the conduction of S2, there must be an input current flowing to support the output current, because the output energy is equal to the input energy (ideal transformer), and because Instantaneous power is equal to the product of voltage and current. From equation (3), we can get

    (Io / nIi) = 1 / D (15)

    After sorting, we can get Ii = IoD / n (16)

    During the conduction of S2, the average load current produces a current on the primary side equal to the input current Ii

    I LP2 D=I i =IoD/n (17)

    or I LP2 =Io/n (18)

    The waveform of the primary winding magnetizing current ILPm is triangular, as shown in Figure 4(f). The load current waveforms shown in equations (14) and (18) are shown in Figure 4(g) and Figure 4(h) respectively, while the resultant primary current waveform is shown in Figure 4(i). Due to the large inductance of the primary winding, ILP2 basically remains at a constant value during the entire switching cycle, even if S2 is turned off.

    If there is no output current, the average value of the magnetizing current is zero. Therefore, when the transformer is no-load, the primary current has a waveform of positive and negative peak-to-peak amplitudes. To obtain a zero-voltage resonant switch, the peak-to-peak amplitude of the magnetizing current must be greater than twice the current generated in the primary winding by the load current.

    The characteristic of this series power conversion topology is that in the forward conversion circuit, only one magnetic component is used, and the magnetic component plays two roles: one is as an inductor in the circuit, and the other is as an isolation transformer. Another similar circuit is shown in Figure 7.

    The circuit structure and working conditions are basically the same as Figure 3. Cp can only be connected to the primary winding when S1 is turned on. The waveform produced by the circuit of Figure 7 is shown in Figure 8. Its working status is shown in Figure 9 and Figure 10 respectively. In Figure 9, S2 is turned on, which increases the current in the primary winding, and the output current is completely provided by the capacitor CS. In Figure 10, S1 and S3 are turned on, and the voltage Ucp on CP (which is formed by the continuous charging and discharging of Cp when S1 and S3 are disconnected) is added to the primary winding of the transformer.

    In steady state, the voltage across the primary inductor averages zero during a switching cycle.

    UiDT＋(-nUo)(1-D)T=0 (19)

    nUo(D-1)+UiD=0 (20)

    nUo / Ui=D / 1-D (21)

    Its waveform is shown in Figure 8(b). From equations (9) and (21), we can get

    Ics=Io·(nUo/Ui) (22)

    The Ics waveform is shown in Figure 8(c). The output current Io waveform is shown in Figure 8(d), while the secondary current Is waveform is shown in Figure 8(e).

    During the conduction period of S2, the magnetizing current

    iLPm(t)=ILPm(t0)+Ui/Lp (23)

    Peak-peak value of magnetizing current:

    (ILPm)pp=(Ui / Lp)DT (24)

    Similarly, during the conduction period of S1 and S3, the peak-to-peak amplitude of the magnetizing current is:

    (ILPm)pp=nUo / Lp(1-D)T (25)

    The waveform of equation (25) is shown in Figure 8(f).

    When the current represented by equation (7) is reflected to the primary side of the transformer, equation (14) is derived.

    During the conduction period of S2, the current generated by the load current on the primary side can be derived from equation (21)

    Io / nIi = 1-D / D (27)

    After sorting, we can get Ii = IoD / n (1-D) ( 28)

    During the entire switching cycle, the current generated on the primary side by the load current during S2 conduction is equal to the input current Ii

    ILP2D=Ii=IoD / n(1-D) (29)

    or ILP2=Io / n(1-D) (30)

    The waveform of equation (30) is shown in Figure 8(g).

    The primary winding magnetizing current ILPm has a triangular waveform, as shown in Figure 8(f). The resultant primary current waveform is shown in Figure 8(h).

    When the output current is zero, as in the case of a forward converter, there is only magnetizing current in the primary winding and its average value is zero. The difference between the circuit in Figure 7 and the circuit in Figure 3 is that in the circuit in Figure 3, there is no magnetizing current in the primary winding during the turn-off period of S2, while in the circuit in Figure 7, CP will still provide a certain magnetizing current even during the turn-off period of S2.

4 Conclusion

    Since the circuit in Figure 3 uses switch S1 as an active clamping/recovery device, the circuit has the following advantages:

    (1) In order to restore the transformer, there is no need to add a recovery winding or a lossy clamping device.

    (2) The duty cycle is relatively high, allowing a wide input voltage range, or using a higher turns ratio.

    (3) Due to the high turns ratio, the current stress on the primary side and the voltage stress on the secondary side can be greatly reduced.

    (4) The energy stored in the parasitic components is transmitted to the resonant tank circuit components and circulated. As a result, the circuit efficiency is improved and the noise is reduced.

    (5) Since the switching voltage is clamped to a controllable level, device stress is reduced, and low-cost switching devices can be used.

    (6) Zero-voltage switching (ZVS) can be implemented, allowing it to operate at higher frequencies and obtain higher efficiency.

    (7) Within the entire input voltage variation range, the voltage stress on the switching tube is quite constant, which provides designers with room for comprehensive considerations. In other single-ended circuits, since the switching voltage stress is proportional to the input voltage, this advantage is not available.

    (8) Due to the use of this active clamping technology, it is possible to use synchronous switching on the secondary side to improve the transformer waveform.