Analog Front-End Design Considerations for RTD Ratiometric Temperature Measurements

Many system designers use Σ-Δ ADCs and RTDs (resistance temperature detectors) for temperature measurement but have difficulty achieving the high performance specified in the ADC data sheet. For example, some designers may only be able to obtain 12 to 13 noise-free bits from a 16-bit to 18-bit ADC. The front-end techniques described in this article enable designers to obtain more than 16 noise-free bits in their system designs.

There are advantages to using an RTD in a ratiometric measurement because it eliminates error sources such as the accuracy and drift of the excitation current source. Below is a typical circuit for a 4-wire RTD ratiometric measurement. The advantage of the 4-wire configuration is that the error introduced by the lead resistance is eliminated.

Figure 1. 4-wire RTD ratiometric measurement circuit.

We can derive the following two formulas from the above circuit:

When the ADC operates in bipolar differential mode, the general expression for calculating the RTD resistance (RRTD) is as follows:

in:

CodeRTD is the ADC code.

CodeADC_Fullscale is the ADC full-scale code.

The measured resistance value of the RTD is theoretically related only to the accuracy and drift of the reference resistor. Typically, RREF is a precise, low-drift resistor with an accuracy of 0.1%.

When engineers design products with this type of circuit, they add some resistors and capacitors before the analog input and external reference pins to get low-pass filtering and overvoltage protection as shown in Figure 2. In this article, we will show what factors should be considered when choosing appropriate resistors and capacitors to get better noise performance.

Figure 2. Typical 4-wire RTD ratiometric measurement circuit.

As can be seen in Figure 2, R1, R2, C1, C2, and C3 act as a first-order low-pass RC filter that provides attenuation for differential and common-mode voltage signals . The values ​​of R1 and R2 should be the same, and the values ​​of C1 and C2 are also selected to be the same. Similarly, R3, R4, C4, C5, and C6 act as a low-pass filter for the reference path.

Common Mode Low Pass RC Filter

Figure 3 shows the common-mode low-pass filter equivalent circuit.

Figure 3. Common-mode low-pass filter.

Because the common-mode voltage at point a is equal to the voltage at point b, no current flows through C3. Therefore, the common-mode cutoff frequency can be expressed as:

Differential Mode Low Pass RC Filter

To better understand the low-pass RC filter cutoff frequency for differential signals, the C3 capacitor in Figure 4 can be viewed as two separate capacitors in Figure 5: Ca and Cb.

Figure 4. Differential mode low-pass filter.

Figure 5. Differential-mode low-pass filter equivalent circuit.

In Figure 5, the differential mode cutoff frequency is:

Typically, the value of C3 is 10 times the value of Ccm. This is to reduce the effect of the inconsistency between C1 and C2. For example, as shown in Figure 6, when using the analog front-end design in ADI Circuit Note CN-0381, the cutoff frequency for differential signals is approximately 800 Hz and the cutoff frequency for common-mode signals is approximately 16 kHz.

Figure 6. Analog input configuration for RTD measurements using the AD7124.

Resistor and capacitor considerations

In addition to being part of the low-pass filter, R1 and R2 also provide overvoltage protection. If a 3 kΩ resistor is used in front of the AD7124-4 AIN pin in Figure 6, protection against up to 30 V wiring errors is possible. Using larger resistors in front of the AIN pin is not recommended for two reasons. First, they will generate more thermal noise. Second, the AIN pin has input currents that will flow through these resistors and introduce errors. These input currents are not constant in magnitude, and unmatched input currents will generate noise, and the noise will increase with larger resistor values.

The resistor and capacitor values ​​are critical in determining the performance of the final circuit. Designers need to understand their site requirements and calculate the resistor and capacitor values ​​based on the above formulas. For ADI Σ-Δ ADC devices and precision analog microcontrollers with integrated excitation current sources, it is recommended to use the same resistor and capacitor values ​​before the AIN and reference voltage pins. This design ensures that the analog input voltage is always proportional to the reference voltage, and any error in the analog input voltage caused by temperature drift and noise of the excitation current can be compensated by the change of the reference voltage.

ADuCM360 Noise Performance Using Ratiometric Measurements

The ADuCM360 is a fully integrated 3.9 kSPS, 24-bit data acquisition system that integrates dual high-performance multi-channel Σ-Δ ADCs, a 32-bit ARM® Cortex®-M3 processor, and Flash/EE memory on a single chip. It also integrates a programmable gain instrumentation amplifier, a precision bandgap reference, a programmable excitation current source, a flexible multiplexer, and many other features. It can be directly connected to resistive temperature sensors.

When using ADuCM360 for RTD measurement, the REF– pin is usually grounded, so R4 and C5 in Figure 2 have no current flowing through them and can be removed. C4 and C6 are connected in parallel. Since C4 is much smaller than C6, it can be ignored. Finally, a simple analog front-end circuit is obtained, as shown in Figure 7.

Figure 7. ADuCM360 analog front-end circuit for RTD measurement.

Table 1 lists the noise levels with matched and unmatched filters in front of the analog and reference input paths. A 100 Ω precision resistor is used in place of the RRTD to measure the noise voltage at the ADC input pins. The value of RRef is 5.62 kΩ.

Table 1. Noise Test Results

From Table 1 we can see that when using a matched analog front-end circuit with R1 and R2 having the same values ​​as R3, the noise is reduced by approximately 0.1 μV to 0.3 μV compared to the unmatched circuit, which means the number of ADC noise-free bits increases by approximately 0.25 bits to 16.2 bits with an ADC PGA gain of 16.

in conclusion

By following the considerations presented in this article, using matched RC filter circuits and selecting the appropriate resistor and capacitor values ​​based on site requirements, the best results can be achieved using RTDs in ratiometric measurement applications.

References

CN-0381 Circuit Note. "Fully Integrated 4-Wire RTD Measurement System Using a Low Power, Precision, 24-Bit Σ-Δ ADC." Analog Devices, Inc.

CN-0267 Circuit Note. "Complete 4 mA to 20 mA Loop-Powered Field Instrument with HART Interface." Analog Devices, Inc.